Radio frequency (RF) power amplifier and RF power amplifier apparatus

ABSTRACT

An RF power amplifier has a final-stage amplifier stage which generates an RF transmit output signal, a signal detector which detects an RF transmit output level, a first detector, a second detector and a control circuit. The final-stage amplifier stage includes a transistor and a load element and performs saturation type nonlinear amplification and non-saturation type linear amplification. The first detector and the control circuit maintain the RF transmit output signal approximately constant with respect to a variation in load at an antenna at the saturation type nonlinear amplification. The second detector and the control circuit reduce an increase in the output voltage of the final stage transistor with respect to an overload state of the antenna at the non-saturation type linear amplification.

CROSS-REFERENCE TO RELATED APPLICATIONS

The disclosure of Japanese Patent Application No. 2008-131421 filed onMay 20, 2008 including the specification, drawings and abstract isincorporated herein by reference in its entirety.

BACKGROUND OF THE INVENTION

The present invention relates to an RF power amplifier and an RF poweramplifier apparatus for RF transmission, each of which is mounted in acommunication terminal device like a cellular phone terminal thatcommunicates with a base station. Further, the present inventionparticularly relates to a technique beneficial to realize two functionsof a non-saturation type linear amplifier and a saturation typenonlinear amplifier by one RF power amplifier.

The operation of a high-frequency power amplifier included in a cellularphone terminal has a saturation operation in GSM of a basic mode inwhich only phase modulation is used. EDGE that uses amplitude modulationalong with phase modulation has linear operation at an operating pointwhere several dB are backed off from a saturation operating point ofGSM. Even in the case of WCDMA and cdma-1x that also use amplitudemodulation along with phase modulation, the operation of thehigh-frequency power amplifier has linear operation.

At a high-frequency circuit portion of the cellular phone terminalcorresponding to each of GSM and EDGE, an antenna switch is disposedbetween the high-frequency power amplifier and an antenna. The antennaswitch performs the function of switching between transmission andreception slots of a TDMA (Time Division Multiple Access) system.

At a high-frequency circuit portion of a cellular phone terminalcorresponding to each of WCDMA and cdma-1x, a duplexer is disposedbetween a high-frequency power amplifier and its corresponding antenna.The duplexer performs the function of processing in parallel,transmission of an RF transmit signal of a low RF frequency of a CDMA(Code-Division Multiple Access) system and reception of an RF receivesignal of a high RF frequency thereof. Further, in WCDMA and cdma-1x orthe like, an isolator is disposed between the high-frequency poweramplifier and the duplexer to avoid the influence of a variation in loadat the antenna over the high-frequency power amplifier. Since it ishowever difficult for the isolator to be integrated into a structure atwhich the high-frequency power amplifier is manufactured, it becomes alarge and expensive part.

Ubiquitous coverage corresponding to the capability of a communicationterminal device such as a cellular phone terminal that wirelesscommunications are carried out in any place in the world is not real inthese days, but now proceeding into development.

According to a non-patent document 1 (Earl McCune, “High-Efficiency,Multi-Mode, Multi-Band Terminal Power Amplifiers”, IEEE microwavemagazine, March 2005, PP. 44-55), these mobile systems respectivelyinclude cells of GSM, GPRS, EDGE and WCDMA, and networks of, forexample, IEEE 802.11-b, -a and -g or the like, such as personal areanetworks such as Bluetooth and ZigBee. The characteristics of thesesystems extend to signals for a constant envelope and a change inenvelope, multiplex time division and code division and a widercombination of transmit output power from high (few watts) to low (microwatts). As a result, there has been a growing demand for amultimode-applied RF power amplifier.

A self-evident approach to the multimode is to apply a linear circuittechnique in order to support an envelope change signal. This approachhowever causes a basic contradiction in the circuit design of a poweramplifier. As is well known, the maximum efficiency of the poweramplifier is obtained by allowing the power amplifier to perform asaturation operation (nonlinear operation at which waveform clipsoccur). Since the power amplifier that performs the saturation operationoperates as a limiter where an input signal is of an envelope changesignal, serious signal distortion occurs. Thus, the power amplifierneeds to perform a non-saturation linear operation in order to reproducethe envelope change signal faithfully. To this end, the concept ofoutput back-off is introduced and the power amplifier is limited to thelinear operation at which the peak output power of the power amplifierthat performs the non-saturation linear operation is lower than themaximum (saturated) output power. Since, however, the output back-offbecomes difficult in design, two discrete power amplifiers correspondingto a non-saturation type linear amplifier and a saturation typenonlinear amplifier have been developed.

Incidentally, GSM is an abbreviation of Global System for MobileCommunication. GPRS is an abbreviation of General Packet Radio Service.Further, EDGE is an abbreviation of Enhanced Data for GSM Evolution(Enhanced Data for GPRS). WCDMA is an abbreviation of Wideband CodeDivision Multiple Access.

On the other hand, an RF power amplifier module that transmits aquadband including frequency bands of GSM850, GSM900, DCS1800 andPCS1900 has been described in non-patent document 2 (Shuyun Zhang et al,“A Novel Power-Amplifier Module for Quad-Band Wireless HandsetApplications”, IEEE TRANSACTIONS ON MICROWAVE THEORY AND TECHNIQUES,VOL. 51, No. 11, NOVEMBER 2003, PP. 2203-2210). Incidentally, DCS is anabbreviation of a Digital Cellular System and PCS is an abbreviation ofa Personal Communication System. The RF power amplifier module includesa first power amplifier which amplifies a first RF transmit input signalhaving a first frequency band of GSM850 and GSM900, and a second poweramplifier which amplifies a second RF transmit input signal having asecond frequency band of DCS1800 and PCS1900.

In communications of GSM850, GSM900, DCS1800 and PCS1900, a TDMA systemhas been adopted which is capable of setting a plurality of time slotsto any of an idle state, a reception operation from a base station and atransmission operation to the base station, respectively, on atime-sharing basis. Incidentally, TDMA is an abbreviation ofTime-Division Multiple Access. As one TDMA system, there is known a GSMsystem that uses only phase modulation. There is also known a systemthat improves a communication data transfer rate as compared with theGSM system. As an improvement system, an EDGE system that uses amplitudemodulation along with the phase modulation has recently been brought toattention.

On the other hand, a WCDMA system that has improved a communication datatransfer rate by using amplitude modulation along with phase modulationhas also been a focus of attention in recent years in a manner similarto the EDGE system. In this WCDMA system, there has been adopted afrequency-division CDMA system other than the TDMA system, which usesfrequencies of 2110 MHz to 2170 MHz for a reception operation from abase station and uses frequencies of 1920 MHz to 1980 MHz for atransmission operation to the base station. Incidentally, CDMA is anabbreviation of Code Division Multiple Access.

A non-patent document 3 (Gary Hau et al, “High Efficiency, Wide DynamicRange Variable Gain and Power Amplifier MMICs for Wide-Band CDMAHandsets”, IEEE MICROWAVE AND WIRELESS COMPONENTS LETTERS, VOL. 11, No.1, JANUARY 2001, PP. 13-15) has described that since a wider controlrange and high linearity are necessary for power control of an RF poweramplifier of the WCDMA system, a programmable gain amplifier using avariable attenuator is coupled to the input of the RF power amplifier.

On the other hand, power control of an RF power amplifier by a closedloop and source voltage control has been described in a non-patentdocument 4 (Angelo Scuderi et al, “A VSWR-Protected Silicon Bipolar RFPower Amplifier With Soft-Slop Power Control”, IEEE JOURNAL OFSOLID-STATE CIRCUITS, VOL. 40, No. 3, MARCH 2005, PP. 611-621). In thepower control by the closed loop, RF power output of the amplifier issensed using a directional coupler and detected by a diode. A detectedvoltage is compared with a reference voltage by an error amplifier. Theoutput of the error amplifier drives a gain control terminal of thepower amplifier to equally control the detected voltage and thereference voltage by the closed loop. The power control is realized by achange in the reference voltage. In the source voltage control of the RFpower amplifier, which controls the output power thereof, a linearregulator configured by a power PMOS transistor and an operationalamplifier is used, and a collector voltage of the RF power amplifier islinearly changed by a control terminal of the operational amplifier. Theoutput amplitude at which power is obtained is limited by reducing thecollector voltage.

Closed-loop collector peak voltage control for coping with a breakdownbased on a high voltage peak at the collector of a final stage in astate of a high-load voltage standing wave ratio (VSWR) caused by loadmismatching has been described in the non-patent document 4. Thiscontrol is configured by an AC sense circuit/envelope detector thatdetect a peak voltage at an output collector node, and an erroramplifier which changes circuit gain thereby to clamp the peak voltageto a specific threshold voltage. The error amplifier controls a biascurrent of a drive stage for driving the final stage.

A parallel power amplifier that realizes low distortion and highefficiency at a wide range of load impedance without using an isolatorhas been described in a non-patent document 5 (Hikaru Ikeda et al, “ALow Distortion and High Efficiency Parallel-Operation Power AmplifierCombined in Different Phases in Wide Range of Load Impedance”, 1996 IEEEMTT-S Digest, pp. 535-538). The parallel power amplifier has a pluralityof amplification paths. A signal inputted to one input terminal issupplied to the inputs of the amplification paths by a hybrid divider.Each of the amplification paths includes an amplifier and a phaseshifter. The phase shifters are disposed on their correspondingamplification paths in such a manner that the phases of operations ofthe amplifiers differ from one another at the amplification paths.Plural outputs of the amplification paths are coupled to a single outputby a hybrid coupler. The non-patent document 5 has described that at a3:1 VSWR (Voltage Standing-Wave Ratio) or less in which reflectivity Γis equivalent to 0.5, a distortion of −45 dBc or less, an efficiency of45% or more and a gain of 9.8 dB or more have been obtained.Incidentally, VSWR is an abbreviation of Voltage Standing-Wave Ratio.VSWR is determined by VSWR=(1+Γ)/(1−Γ) in accordance with reflectivityΓ.

A balanced amplifier similar to the parallel power amplifier describedin the non-patent document 5 has been described in a non-patent document6 (Giuseppe Berrtta et al, “A balanced CDMA SiGe HBT Load InsensitivePower Amplifier”, 2006 IEEE Radio and Wireless Symposium 17-19 Jan.2006, PP. 523-526) to adapt to the mismatching of a load due to theomission of the isolator. The balanced amplifier includes an inputhybrid coupler, two RF power amplifiers, two matching circuits, anoutput hybrid coupler and two 50Ω terminal resistors. The supply of anRF input signal to input-side two terminals of the input hybrid couplerand the coupling of the 50Ω input terminal resistors thereto are carriedout. Two input terminals of the two RF power amplifiers are coupled totheir corresponding two output-side terminals of the input hybridcoupler. Two input terminals of the two matching circuits are coupled totheir corresponding two output terminals of the two RF power amplifiers.Two input-side terminals of the output hybrid coupler are coupled totheir corresponding two output terminals of the two matching circuits.The coupling of the 50Ω output terminal resistors to their correspondingtwo output-side terminals of the output hybrid coupler and the couplingof an antenna thereto are carried out. The parallel power amplifierdemonstrates a satisfactory input/output return loss and hasinsensitivity satisfactory for variations in a load.

SUMMARY OF THE INVENTION

Prior to the present invention, the present inventors have been involvedin the development of an RF power amplifier module that transmits fivefrequency bands of GSM850, GSM900, DCS1800, PCS1900 and WCDMA2100.

In the development of the RF power amplifier module prior to the presentinvention, miniaturization of the RF power amplifier module was requiredto achieve a further scale-down of a cellular phone terminal thatsupports multimode communications of GSM, EDGE and WCDMA. It wastherefore necessary to realize two functions of a non-saturation typelinear amplifier and a saturation type nonlinear amplifier by one RFpower amplifier. As described in the non-patent document 1, the twodiscrete power amplifiers of the non-saturation type linear amplifierand the saturation type nonlinear amplifier have heretofore beendeveloped and used due to the design difficulty. However, the use of thetwo discrete power amplifiers will no longer be allowed to achievefurther scaling down of the cellular phone terminal that supports themultimode communications.

In GSM communications at GSM850, GSM900, DCS1800 and PCS1900, a constantenvelope signal called “GMSK” is used. Incidentally, GMSK is anabbreviation of Gaussian minimum shift keying. On the other hand, anenvelope change signal called “3π/8-8PSK” is used in EDGE communicationsat GSM850, GSM900, DCS1800 and PCS1900. Incidentally, 3π/8-8PSK is anabbreviation of 8-phase-shift keying with a 3π/8 phase shift addedcumulatively to each symbol”. In communications at WCDMA, an envelopechange signal called “HPSK” is used. Incidentally, HPSK is anabbreviation of Hybrid phase shift keying.

Thus, the present inventors have achieved the idea that one RF poweramplifier is commonly used in the saturation type nonlinear amplifierand the non-saturation type linear amplifier, and the operations of afirst operation mode for saturation type nonlinear amplification and asecond operation mode for non-saturation type linear amplification areexecuted.

Namely, upon transmission of the GMSK constant envelope signal in theGSM communications, the operation mode of one RF power amplifier is setto the first operation mode for the saturation type nonlinearamplification thereby to make it possible to realize a high-efficientoperation. Upon transmission of the envelope change signals at 3π/8-8PSKin the EDGE communications and HPSK in the WCDMA communications, theoperation mode of one RF power amplifier is set to the second operationmode for the non-saturation type linear amplifier thereby to make itpossible to reduce signal distortion of the envelope change signal.

Prior to the present invention, the present inventors have discussedthat the balanced RF power amplifier excellent in load variationsuppression characteristic described in each of the non-patent documents5 and 6 is applied to the RF power amplifier module that transmits thefive frequency bands of GSM850, GSM900, DCS1800, PCS1900 and WCDMA2100.The balanced RF power amplifier adopts such a configuration that RFinput signals different by 90° in phase are supplied to the two RF poweramplifiers by the input hybrid coupler, and their phases are furtherrotated 90° by the output hybrid coupler based on the outputs of the twomatching circuits. Even if the mismatch of impedance between the outputof each power amplifier and the antenna occurs in the balanced RF poweramplifier, the whole ACPR (adjacent channel leakage power ratio) of thebalanced power amplifier can be rendered satisfactory. This is becausethe impedance transformation of the power amplifier for one of the twoamplification paths becomes an inductive rotation on the Smith chart,and the impedance transformation of the power amplifier for the otherthereof becomes a capacitive rotation on the Smith chart. As a result,when one impedance is brought to high impedance, the other impedancebecomes low impedance, thereby making it possible to correct distortionof a combined signal.

When, however, errors from 90° occur in the phase rotations between theinput hybrid coupler and the output hybrid coupler in the balanced RFpower amplifier, losses occur in the matching circuits, thus leading toa reduction in efficiency. Since the input and output hybrid couplersare respectively comprised of circuits complex and large in the numberof elements and include resistors along with a plurality of capacitorsand a plurality of inductors, the problem that the loss in each resistoroccurs and high efficiency is difficult has been revealed.

Thus, the present inventors have taken one RF power amplifier which iscommonly used in the saturation type nonlinear amplifier and thenon-saturation type linear amplifier based on the above idea and whoseoperation is switched to the first operation mode for the saturationtype nonlinear amplification and the second operation mode for thenon-saturation type linear amplification, as the general signal typewithout being taken as the balanced type.

Thus, it has been revealed that one RF power amplifier which is commonlyused in the saturation type nonlinear amplifier and non-saturation typelinear amplifier of the general single type and whose operation isswitched to the first operation mode and the second operation mode,needs to cope with a variation in load.

Further, it has also been revealed that there is a need to reduce theadjacent channel leakage power ratio (ACPR) of the single typecommonly-used RF power amplifier upon adaptation to the load variationin the second operation mode for the non-saturation type linearamplification. Incidentally, ACPR is an abbreviation of Adjacent ChannelPower leakage Ratio.

It has been demonstrated by discussions of the present inventors thatwhen the load of the antenna is extremely small, there is a need tosuppress the wasting of a battery for the cellular phone terminal andsuppress an operating current flowing through a transistor of afinal-stage amplifier stage in the single type commonly-used RF poweramplifier upon GSM communications large particularly in RF transmitpower. Namely, it has been revealed by the present inventors that if theoperating current is not suppressed in such a case, then the battery iswasted abruptly, and characteristic degradation of the transistor of thefinal-stage amplifier stage occurs and a current breakdown occurs.

The present invention has been made as a result of the discussions ofthe present inventors prior to the present invention referred to above.

Thus, an object of the present invention is to cope with a loadvariation and an overload state in one RF power amplifier which iscommonly used in general single type saturation-type nonlinear andnon-saturation type linear amplifiers and which executes a firstoperation mode for saturation type nonlinear amplification and a secondoperation mode for non-saturation type linear amplification.

Another object of the present invention is to reduce an adjacent channelleakage power ratio (ACPR) in an overload state of a single typecommonly-used RF power amplifier upon adaptation to a load variation ina second operation mode for non-saturation type linear amplification.

A further object of the present invention is to suppress an operatingcurrent that flows through a transistor of a final-stage amplifier stageof a single type commonly-used RF power amplifier where the load of anantenna is extremely small, upon GSM communications large particularlyin RF transmit power.

The above and other objects and novel features of the present inventionwill become apparent from the description of the present specificationand the accompanying drawings.

A representative embodiment of the present invention disclosed in thepresent application will be described in brief as follows:

A typical RF power amplifier (100) of the present invention has afinal-stage amplifier stage (11) which amplifies an RF signal togenerate an RF transmit output signal (P_(OUT)) supplied to an antennaof a communication terminal, and a signal detector (13) which detectsthe level of the RF transmit output signal. The final-stage amplifierstage includes a final stage transistor (Qn2) and a final stage loadelement (L2) and executes a first operation mode for saturation typenonlinear amplification and a second operation mode for non-saturationtype linear amplification.

The RF power amplifier (100) further includes a first detector (14), asecond detector (17) and a controller or control circuit (15, 16 and18). The first detector (14) is supplied with an RF detection signalfrom the signal detector (13). The second detector (17) is supplied withan output voltage (Vds) of the final stage transistor.

First feedback control by the first detector and the control circuitreduces a variation in the RF transmit output signal (P_(OUT)) generatedfrom the final stage transistor responsive to a variation in load at theantenna in the first operation mode.

Second feedback control by the second detector and the control circuitreduces an increase in the output voltage (Vds) of the final stagetransistor responsive to an overload state of the antenna in the secondoperation mode (refer to FIG. 1).

Advantageous effects obtained by a representative embodiment of thepresent invention disclosed in the present application will be explainedin brief as follows: According to the present invention, it is possibleto cope with a load variation and an overload state in one RF poweramplifier which is commonly used in general single type saturation-typenonlinear and non-saturation type linear amplifiers and which executes afirst operation mode for saturation type nonlinear amplification and asecond operation mode for non-saturation type linear amplification.

According to the present invention as well, it is possible to reduce anadjacent channel leakage power ratio (ACPR) in an overload state of asingle type commonly-used RF power amplifier upon adaptation to a loadvariation in a second operation mode for non-saturation type linearamplification.

BRIEF DESCRIPTION OF THE DRAWINGS

FIG. 1 is a diagram showing a configuration of a cellular phone terminalincluding an RF power amplifier according to an embodiment of thepresent invention and a baseband digital signal processing/RF analogsignal processing unit for supplying an RF transmit signal to the RFpower amplifier;

FIG. 2 is a diagram for describing the operation of the RF poweramplifier of FIG. 1 when its load impedance is changed;

FIG. 3 is a diagram for describing the dependence of a drain-sourcevoltage of a transistor of a final-stage amplifier stage when the loadimpedance of the RF power amplifier shown in FIG. 1 is brought to astate of being a value smaller than a proper value, a state of being theproper value and a state of being a value larger than the proper value;

FIG. 4 is a diagram for describing the dependence of an RF transmitoutput signal of the RF power amplifier when the load impedance of theRF power amplifier shown in FIG. 1 is brought to a state of being avalue smaller than a proper value, a state of being the proper value anda state of being a value larger than the proper value;

FIG. 5 is a diagram for describing the dependence between a saturationstart of an RF transmit output signal of the RF power amplifier and asudden increase in adjacent channel leakage power ratio when the loadimpedance of the RF power amplifier shown in FIG. 1 is brought to astate of being a value smaller than a proper value, a state of being theproper value and a state of being a value larger than the proper value;

FIG. 6 is a diagram showing input/output characteristics of first andsecond detectors of the RF power amplifier shown in FIG. 1;

FIG. 7 is a diagram illustrating configurations of the first and seconddetectors of the RF power amplifier shown in FIG. 1;

FIG. 8 is a diagram showing one example illustrative of an outputmatching circuit and a directional coupler both coupled to a drainoutput electrode of a common-source N channel MOS transistor of thefinal-stage amplifier stage of the RF power amplifier shown in FIG. 1;

FIG. 9 is a diagram illustrating a configuration of an RF poweramplifier in which an adjustment circuit mounted to the RF poweramplifier shown in FIG. 1 and having coefficients set to constantsgreater than or equal to 1 is replaced with an adjustment circuit havingnon-linear characteristics;

FIG. 10 is a diagram showing a configuration of an RF power amplifier towhich an overcurrent protection circuit for adapting to a load shortcircuit when a large RF transmit signal used in GSM communications istransmitted to the RF power amplifier shown in FIG. 1;

FIG. 11 is a diagram illustrating a configuration of an RF poweramplifier wherein in the RF power amplifier shown in FIG. 1, firstfeedback from the output of the first detector to a controller of asignal processing unit and second feedback from the output of the seconddetector to the controller of the signal processing unit are renderedindependent; and

FIG. 12 is a diagram showing a configuration of a cellular phone inwhich an RF power amplifier according to an embodiment of the presentinvention and a signal processing unit are provided as a dual-bandconfiguration that covers low and high band frequencies, and whichincludes a DC/DC converter, an antenna switch, duplexers or the like andan antenna.

DETAILED DESCRIPTION OF THE PREFERRED EMBODIMENTS RepresentativeEmbodiments

A summary of representative embodiments of the invention disclosed inthe present application will first be explained. Reference numerals inthe figures which are referred to with parentheses in the outlinedescription of the typical embodiments just exemplify ones contained inconcepts of components to which the reference numerals are attachedrespectively.

[1] An RF power amplifier (100) according to a typical embodiment of thepresent invention includes a final-stage amplifier stage (11) whichamplifies an RF signal to generate an RF transmit output signal(P_(OUT)) supplied to an antenna of a communication terminal, and asignal detector (13) which detects the level of the RF transmit outputsignal.

The final-stage amplifier stage includes a final stage transistor (Qn2)which generates the RF transmit output signal at its output electrode,and a final stage load element (L2) coupled between the output electrodeof the final stage transistor and a source or power supply voltage(Vdd). The final-stage amplifier stage executes a first operation modefor saturation type nonlinear amplification and a second operation modefor non-saturation type linear amplification.

An input terminal of the signal detector (13) is coupled to the outputelectrode of the final stage transistor to thereby generate an RFdetection signal responsive to the level of the RF transmit outputsignal from an output terminal of the signal detector.

The RF power amplifier (100) further includes a first detector (14), asecond detector (17) and a controller or control circuit (15, 16 and18).

An input terminal of the first detector (14) is supplied with the RFdetection signal from the output terminal of the signal detector (13) tothereby generate a first detection signal from an output terminal of thefirst detector (14).

An input terminal of the second detector (17) is coupled to the outputelectrode of the final stage transistor to thereby generate a seconddetection signal from an output terminal of the second detector (17).

The first detection signal generated from the output terminal of thefirst detector (14) contains a first detection component responsive to avariation in the level of the RF transmit output signal due to avariation in load at the antenna when the RF power amplifier (100)operates in the first operation mode for the saturation type nonlinearamplification.

The second detection signal generated from the output terminal of thesecond detector (17) contains a second detection component responsive toan increase in output voltage (Vds) of the output electrode of the finalstage transistor (Qn2) due to an overload state of the antenna when theRF power amplifier (100) operates in the second operation mode for thenon-saturation type linear amplification.

The first detector (14) and the second detector (17) respectively have afirst input threshold voltage (Vth1) and a second input thresholdvoltage (Vth2). The level of the second input threshold voltage is sethigher than the level of the first input threshold voltage.

The first detection signal generated from the output terminal of thefirst detector and the second detection signal generated from the outputterminal of the second detector are supplied to the control circuit (15,16 and 18). The control circuit controls the levels of the RF transmitoutput signal and the output voltage at the output electrode of thefinal stage transistor.

First feedback control by the first detector and the control circuitreduces a variation in the RF transmit output signal (P_(OUT)) generatedfrom the output electrode of the final stage transistor (Qn2) responsiveto the load variation of the antenna upon the operation in the firstoperation mode.

Second feedback control by the second detector and the control circuitreduces the increase in the output voltage (Vds) of the output electrodeof the final stage transistor (Qn2) responsive to the overload state ofthe antenna upon the operation in the second operation mode (refer toFIG. 1).

According to the embodiment, the RF transmit output signal (P_(OUT))generated from the output electrode of the final stage transistor (Qn2)is maintained approximately constant by the first feedback control evenif the antenna load variation has occurred in the first operation mode.Even if the overload state of the antenna has occurred in the secondoperation mode, the increase in the output voltage (Vds) of the outputelectrode of the final stage transistor (Qn2) is reduced by the secondfeedback control. Thus, one RF power amplifier which executes the firstoperation mode for the saturation type nonlinear amplification and thesecond operation mode for the non-saturation type linear amplificationboth commonly used in general single saturation-type nonlinear andnonsaturation-type linear amplifiers, can adapt to the load state andthe overload state.

Further, according to the embodiment, since the increase in the outputvoltage (Vds) of the final stage transistor (Qn2) is reduced where theoverload state of the antenna occurs in the second operation mode, anincrease in adjacent channel leakage power ratio (ACPR) can be reduced.

In a preferred embodiment, the RF signal is of a constant envelopesignal used in GSM communications in the first operation mode for thesaturation type nonlinear amplification. In the second operation modefor the non-saturation type linear amplification, the RF signal is of anenvelope change signal used in either one of EDGE and WCDMAcommunications.

In another preferred embodiment, the signal detector is of a directionalcoupler (13) having a main line and a sub-line disposed close to eachother approximately in parallel.

In yet another preferred embodiment, an output matching circuit iscoupled to the output electrode of the final stage transistor (Qn2). Themain line of the directional coupler is coupled between the outputmatching circuit and the antenna. An RF detection signal responsive tothe level of the RF transmit output signal is generated from thesub-line of the directional coupler.

In a further preferred embodiment, the RF power amplifier (100) includesa first-stage amplifier stage (10).

The RF signal amplified by the final-stage amplifier stage (11) isgenerated from the first-stage amplifier stage (10).

The first-stage amplifier stage (10) includes a first stage transistor(Qn1) which generates the RF signal. A programmable gain amplifier (21)which supplies an RF transmit input signal (P_(IN)) is coupled to acontrol input terminal of the first stage transistor.

The control circuit (15, 16 and 18) controls the programmable gain ofthe programmable gain amplifier (21) to thereby control the levels ofthe RF transmit output signal and the output voltage at the outputelectrode of the final stage transistor.

In one concrete embodiment, the control circuit includes an adder (15)which adds a component of the first detection signal generated from theoutput terminal of the first detector and a component of the seconddetection signal generated from the output terminal of the seconddetector, and a low-pass filter (16) coupled to the output of the adder.

An analog control signal of the control circuit that controls the levelsof the RF transmit output signal and the output voltage at the outputelectrode of the final stage transistor is generated from the output ofthe low-pass filter (16).

In another concrete embodiment, the first stage transistor of thefirst-stage amplifier stage and the final stage transistor of thefinal-stage amplifier stage are either one of an LDMOS transistor and ahetero bipolar transistor respectively.

In the most concrete embodiment, the RF power amplifier (100) is furtherequipped with a current sense circuit (19) and an overcurrent detectioncircuit (18B).

The current sense circuit (19) includes a current sense transistor (Qn3)which has an element size smaller than the final stage transistor (Qn2)of the final-stage amplifier stage (11) and causes a sense currentsmaller than current flowing through the final stage transistor (Qn2) toflow therethrough.

The overcurrent detection circuit (18B) supplies an overcurrentdetection signal to the control circuit (15, 16 and 18) in response tothe sense current that flows through the current sense transistor (Qn3)of the current sense circuit (19).

When the current flowing through the final stage transistor (Qn2) isbrought to an overcurrent state due to the load variation of the antennawhere the RF power amplifier (100) operates in the first operation modefor the saturation type nonlinear amplification, the overcurrentdetection circuit (18B) performs overcurrent protection of the finalstage transistor (refer to FIG. 10).

[2] An RF power amplifier apparatus (100) according to a typicalembodiment of another aspect of the present invention includes a firstRF power amplifier and a second RF power amplifier.

The first RF power amplifier amplifies a first RF transmit input signalhaving a first frequency band ranging from approximately 0.8 GHz toapproximately 1.0 GHz.

The second RF power amplifier amplifies a second RF transmit inputsignal having a second frequency band ranging from approximately 1.7 GHzto approximately 2.0 GHz.

Power amplifiers of the first RF power amplifier and the second RF poweramplifier respectively include final-stage amplifier stages (11_LB, HB)each of which amplifies an RF signal to generate an RF transmit outputsignal supplied to an antenna of a communication terminal, and signaldetectors (13_LB, HB) each of which detects the level of the RF transmitoutput signal.

The final-stage amplifier stage of each of the power amplifiers includesa final stage transistor (Qn2) which generates the RF transmit outputsignal at its output electrode, and a final stage load element (L2)coupled between the output electrode of the final stage transistor and asource or power supply voltage (Vdd).

Each of the power amplifiers executes a first operation mode forsaturation type nonlinear amplification and a second operation mode fornon-saturation type nonlinear amplification.

In the respective power amplifiers, the input terminals of the signaldetectors (13_LB, HB) are respectively coupled to the output electrodesof the final stage transistors to thereby generate RF detection signalseach responsive to the level of the RF transmit output signal from theoutput terminals of the signal detectors.

The power amplifiers further include first detectors (14_LB, HB), seconddetectors (17, 18_LB, HB) and control circuits (15_LB, HB and 16_LB, HB)respectively.

In the respective power amplifiers, the input terminals of the firstdetectors (14_LB, HB) are respectively supplied with the RF detectionsignals from the output terminals of the signal detectors (13_LB, HB) tothereby generate first detection signals from the output terminals ofthe first detectors (14_LB, HB).

In the respective power amplifiers, the input terminals of the seconddetectors (17, 18_LB, HB) are respectively coupled to the outputelectrodes of the final stage transistors to thereby generate seconddetection signals from the output terminals of the second detectors (17,18_LB, HB).

In each power amplifier, the first detection signal generated from theoutput terminal of the first detector contains a first detectioncomponent responsive to a variation in the level of the RF transmitoutput signal due to a load variation of the antenna when the poweramplifier operates in the first operation mode for the saturation typenonlinear amplification.

In each power amplifier, the second detection signal generated from theoutput terminal of the second detector contains a second detectioncomponent responsive to an increase in the output voltage of the outputelectrode of the final stage transistor due to an overload state of theantenna when the power amplifier operates in the second operation modefor the non-saturation type linear amplification.

In the power amplifiers, the first and second detectors have first andsecond input threshold voltages (Vth1) and (Vth2) respectively. Thelevel of the second input threshold voltage is set higher than that ofthe first input threshold voltage.

In each power amplifier, the first detection signal generated from theoutput terminal of the first detector and the second detection signalgenerated from the output terminal of the second detector are suppliedto the control circuit. The control circuit controls the levels of theRF transmit output signal and the output voltage at the output electrodeof the final stage transistor.

In each power amplifier, first feedback control by the first detectorand the control circuit reduces the variation in the RF transmit outputsignal generated from the output electrode of the final stage transistorresponsive to the load variation of the antenna upon its operation inthe first operation mode.

In each power amplifier, second feedback control by the second detectorand the control circuit reduces the increase in the output voltage ofthe output electrode of the final stage transistor responsive to theoverload state of the antenna upon its operation in the second operationmode (refer to FIG. 12).

In a preferred embodiment, the RF signal is of a constant envelopesignal used in GSM communications in the first operation mode for thesaturation type nonlinear amplification. In the second operation modefor the non-saturation type linear amplification, the RF signal is of anenvelope change signal used in either one of EDGE and WCDMAcommunications.

In another preferred embodiment, the signal detector in each poweramplifier is of a directional coupler (13) having a main line and asub-line disposed close to each other approximately in parallel.

In a more preferred embodiment, an output matching circuit is coupled tothe output electrode of the final stage transistor in each poweramplifier. The main line of the directional coupler is coupled betweenthe output matching circuit and the antenna. An RF detection signalresponsive to the level of the RF transmit output signal is generatedfrom the sub-line of the directional coupler.

In a still more preferred embodiment, the RF power amplifiers furtherinclude first-stage amplifier stages (10_LB, HB).

In the power amplifiers, the RF signals amplified by the final-stageamplifier stages (11_LB, HB) are generated from the first-stageamplifier stages (10_LB, HB) respectively.

In the power amplifiers, the first-stage amplifier stages (10_LB, HB)include first stage transistors (Qn1) which generate the RF signals. Aprogrammable gain amplifier (21) which supplies an RF transmit inputsignal (P_(IN)), is coupled to a control input terminal of each of thefirst stage transistors.

In the power amplifiers, the control circuits (15_LB, HB, 16_LB, HB and17, 18_LB, HB) respectively controls the programmable gains of theprogrammable gain amplifiers (21) to thereby control the levels of theRF transmit output signals and the output voltages at the outputelectrodes of the final stage transistors.

In one concrete embodiment, the control circuit of each of the poweramplifiers includes an adder which adds a component of the firstdetection signal generated from the output terminal of the firstdetector and a component of the second detection signal generated fromthe output terminal of the second detector, and a low-pass filtercoupled to the output of the adder.

In each of the power amplifiers, an analog control signal of the controlcircuit that controls the levels of the RF transmit output signal andthe output voltage at the output electrode of the final stage transistoris generated from the output of each of the low-pass filters (16_LB,HB).

In another concrete embodiment, the first stage transistors of thefirst-stage amplifier stages and the final stage transistors of thefinal-stage amplifier stages are respectively either one of an LDMOStransistor and a hetero bipolar transistor.

In the most concrete embodiment, each of the RF power amplifiers isfurther equipped with a current sense circuit (19) and an overcurrentdetection circuit (18B).

In each of the power amplifiers, the current sense circuit (19) includesa current sense transistor (Qn3) which has an element size smaller thanthe final stage transistor (Qn2) of the final-stage amplifier stage (11)and causes a sense current smaller than current flowing through thefinal stage transistor (Qn2) to flow therethrough.

In each of the power amplifiers, the overcurrent detection circuit (18B)supplies an overcurrent detection signal to the control circuit (15, 16and 18) in response to the sense current that flows through the currentsense transistor (Qn3) of the current sense circuit (19).

When the current flowing through the final stage transistor (Qn2) isbrought to an overcurrent state due to the load variation of the antennawhere each of the RF power amplifiers (100) operates in the firstoperation mode for the saturation type nonlinear amplification, theovercurrent detection circuit (18B) performs overcurrent protection ofthe final stage transistor in each of the power amplifiers.

[3] An RF power amplifier (100) according to a typical embodiment of afurther aspect of the present invention has a final-stage amplifierstage (11) which amplifies an RF signal to generate an RF transmitoutput signal (P_(OUT)) supplied to an antenna of a communicationterminal, a control circuit (15, 16 and 18), a current sense circuit(19) and an overcurrent detection circuit (18B).

The final-stage amplifier stage includes a final stage transistor (Qn2)which generates the RF transmit output signal at its output electrode,and a final stage load element (L2) coupled between the output electrodeof the final stage transistor and a source voltage (Vdd).

While the control circuit controls the level of the RF transmit outputsignal of the output electrode of the final stage transistor in thefinal-stage amplifier stage, it executes a first operation mode forsaturation type nonlinear amplification and a second operation mode fornon-saturation type linear amplification.

In the first operation mode for the saturation type nonlinearamplification, the RF signal is of a constant envelope signal used inGSM communications. In the second operation mode for the non-saturationtype linear amplification, the RF signal is of an envelope change signalused in either one of EDGE and WCDMA communications.

The current sense circuit (19) includes a current sense transistor (Qn3)which has an element size smaller than the final stage transistor (Qn2)of the final-stage amplifier stage (11) and causes a sense currentsmaller than current flowing through the final stage transistor (Qn2) toflow therethrough.

The overcurrent detection circuit (18) supplies an overcurrent detectionsignal to the control circuit (15, 16 and 18) in response to the sensecurrent that flows through the current sense transistor (Qn3) of thecurrent sense circuit (19).

The RF power amplifier (100) operates in the first operation mode forthe saturation type nonlinear amplification to thereby transmit the RFtransmit output signal (P_(OUT)) of the constant envelope signal in theGSM communications.

The RF power amplifier (100) operates in the second operation mode forthe non-saturation type linear amplification to thereby transmit the RFtransmit output signal (P_(OUT)) of the envelope change signal in eitherone of the EDGE and WCDMA communications.

When the current flowing through the final stage transistor (Qn2) isbrought to an overcurrent state due to the load variation of the antennaupon transmission of the RF transmit output signal (P_(OUT))) of theconstant envelope signal in the GSM communications by the firstoperation mode of the RF power amplifier, the overcurrent detectioncircuit (18B) performs overcurrent protection of the final stagetransistor (refer to FIG. 10).

According to the embodiment, it is possible to suppress an operatingcurrent flowing through a single type commonly-used RF power amplifierwhere the load of an antenna is extremely small upon GSM communicationslarge in RF transmission power.

In a preferred embodiment, the RF power amplifier (100) further includesan output matching circuit, a directional coupler (13), a first detector(14) and a second detector (17).

The output matching circuit is coupled to the output electrode of thefinal stage transistor in the final-stage amplifier stage.

The directional coupler (13) has a main line and a sub-line disposedclose to each other approximately in parallel. The main line is coupledbetween the output matching circuit and the antenna. An RF detectionsignal responsive to the level of the RF transmit output signal(P_(OUT)) is generated from the sub-line.

An input terminal of the first detector (14) is supplied with the RFdetection signal from the sub-line of the directional coupler (13) tothereby generate a first detection signal from an output terminal of thefirst detector (14).

An input terminal of the second detector (17) is coupled to the outputelectrode of the final stage transistor to thereby generate a seconddetection signal from an output terminal of the second detector (17).

The first detection signal generated from the output terminal of thefirst detector (14) contains a first detection component responsive to avariation in the level of the RF transmit output signal due to avariation in load at the antenna when the RF power amplifier (100)operates in the first operation mode for the saturation type nonlinearamplification.

The second detection signal generated from the output terminal of thesecond detector (17) contains a second detection component responsive toan increase in output voltage (Vds) of the output electrode of the finalstage transistor (Qn2) due to an overload state of the antenna when theRF power amplifier (100) operates in the second operation mode for thenon-saturation type linear amplification.

The first detector (14) and the second detector (17) respectively have afirst input threshold voltage (Vth1) and a second input thresholdvoltage (Vth2). The level of the second input threshold voltage is sethigher than the level of the first input threshold voltage.

The first detection signal generated from the output terminal of thefirst detector and the second detection signal generated from the outputterminal of the second detector are supplied to the control circuit (15,16 and 18). The control circuit controls the levels of the RF transmitoutput signal and the output voltage at the output electrode of thefinal stage transistor.

First feedback control by the first detector and the control circuitreduces a variation in the RF transmit output signal (P_(OUT)) generatedfrom the output electrode of the final stage transistor (Qn2) responsiveto the load variation of the antenna upon the operation in the firstoperation mode.

Second feedback control by the second detector and the control circuitreduces the increase in the output voltage (Vds) of the output electrodeof the final stage transistor (Qn2) responsive to the overload state ofthe antenna upon the operation in the second operation mode (refer toFIG. 10).

In a more preferred embodiment, the first stage transistor of thefirst-stage amplifier stage and the final stage transistor of thefinal-stage amplifier stage are respectively either one of an LDMOStransistor and a hetero bipolar transistor.

Description of Embodiment

Embodiments will next be descried in further detail.

<<Configuration of Cellular Phone Terminal>>

FIG. 1 is a diagram showing a configuration of a cellular phone terminalincluding an RF power amplifier 100 according to an embodiment of thepresent invention and a baseband digital signal processing/RF analogsignal processing unit 200 which supplies an RF transmit signal to theRF power amplifier 100.

The cellular phone terminal shown in FIG. 1 performs GSM, EDGE and WCDMAcommunications with various base stations.

The baseband digital signal processing of the signal processing unit 200generates a digital baseband transmit signal, which is converted to ananalog baseband transmit signal by a D/A converter. The analog basebandtransmit signal is frequency-upconverted to an RF transmit signal by anRF transmit signal processing unit including a transmit voltage controloscillator 20. The RF transmit signal from the transmit voltage controloscillator 20 is supplied to an RF transmit signal input terminal of theRF power amplifier 100 via a programmable gain amplifier 21.

A gain control analog signal for setting the amplification gain of theprogrammable gain amplifier 21 of the signal processing unit 200 issupplied from the RF power amplifier 100 to an A/D converter 22 of thesignal processing unit 200. The A/D converter 22 converts the gaincontrol analog signal to a gain control digital signal. A controller 23sets the amplification gain of the programmable gain amplifier 21 inresponse to the gain control digital signal from the A/D converter 22.

<<RF Power Amplifier that Supports Multimode>>

The RF power amplifier 100 shown in FIG. 1 amplifiers an RF transmitinput signal P_(IN) from the signal processing unit 200 to therebygenerate an RF transmit output signal P_(OUT). The RF transmit outputsignal P_(OUT) is transmitted to a base station through an antenna (notshown) of a cellular phone terminal. Upon amplification of the RFsignal, the RF power amplifier 100 executes a first operation mode forsaturation type nonlinear amplification used for transmitting a GMSKconstant envelope signal in GSM communications, and a second operationmode for non-saturation type linear amplification used for transmittingenvelop change signals of 3π/8-8PSK in EDGE communications and HPSK inWCDMA communications. That is, the RF power amplifier 100 shown in FIG.1 is used in common to the first operation mode for the saturation typenonlinear amplification and the second operation mode for thenon-saturation type linear amplification. One RF power amplifier 100executes the first operation mode and the second operation mode.

<<Configuration of RF Power Amplifier>>

The RF power amplifier 100 shown in FIG. 1 first includes a first-stageamplifier stage 10, a final-stage amplifier stage 11, a bias circuit 12and a directional coupler 13 formed in a silicon semiconductor chip.

The first-stage amplifier stage 10 includes a common-source N channelMOS transistor Qn1 and a choke coil L1 used as a load. The source of thecommon-source N channel MOS transistor Qn1 is coupled to a groundvoltage, the gate thereof is supplied with an RF transmit input signalP_(IN) in any of GSM, EDGE and WCDMA communications, and the drainthereof is supplied with a source or power supply voltage Vdd via thechoke coil L1.

In a manner similar to the first-stage amplifier stage 10, thefinal-stage amplifier stage 11 includes a common-source N channel MOStransistor Qn2 and a choke coil L2 used as a load. The source of thecommon-source N channel MOS transistor Qn2 is coupled to the groundvoltage, the gate thereof is supplied with a drain amplification signalof the common-source N channel MOS transistor Qn1 of the first-stageamplifier stage 10 via a capacitor C1, and the drain thereof is suppliedwith the source voltage Vdd through the choke coil L2. An RF transmitoutput signal P_(OUT) is generated from the drain of the common-source Nchannel MOS transistor Qn2 of the final-stage amplifier stage 11. The RFtransmit output signal P_(OUT) is supplied via an output matchingcircuit (not shown) and the directional coupler 13 to an antenna of acellular phone terminal with the RF power amplifier 100 mounted thereto.Since the amplification gain of the final-stage amplifier stage 11 isset larger than that of the first-stage amplifier stage 10 inparticular, the element size of the common-source N channel MOStransistor Qn2 of the final-stage amplifier stage 11 is set far largerthan the element size of the common-source N channel MOS transistor Qn1of the first-stage amplifier stage 10. This element size is set by agate area based on the product of a gate width of each MOS transistorand a gate length thereof.

The common-source N channel MOS transistor Qn1 of the first-stageamplifier stage 10 and the common-source N channel MOS transistor Qn2 ofthe final-stage amplifier stage 11 are respectively configured by an RFamplifying LDMOS transistor lying in the silicon semiconductor chip.However, these common-source N channel MOS transistors can also bereplaced with a common-emitter NPN type HBT (Hetero junction BipolarTransistor) respectively. When the NPN type HBT is used, the elementsize is set by its emitter area. Incidentally, LDMOS is an abbreviationof Laterally Diffused Metal Oxide Semiconductor. Also HBT is anabbreviation of Hetero junction Bipolar Transistor.

The RF power amplifier 100 includes a bias circuit 12. The bias circuit12 generates a bias voltage Vgs1 supplied to the gate of thecommon-source N channel MOS transistor Qn1 of the first-stage amplifierstage 10, and a bias voltage Vgs2 supplied to the gate of thecommon-source N channel MOS transistor Qn2 of the final-stage amplifierstage 11.

The bias voltages Vgs1 and Vgs2 are set to approximately constant biasvoltages unrelated to variations in source voltage Vdd and ambienttemperature Ta, but can also be made proportional to the level of a rampvoltage Vramp (not shown) from the signal processing unit 200. The rampvoltage Vramp is of a transmit output level instruction signalproportional to the communication distance between the cellular phoneterminal and the base station. The ram voltage Vramp generated at thesignal processing unit 200 in accordance with a signal received from thebase station is supplied to the bias circuit 12, where the levels of thebias voltages Vgs1 and Vgs2 can be determined.

The bias voltages Vgs1 and Vgs2 can also be set as bias voltagesdifferent in the first operation mode for the saturation type nonlinearamplifier and the second operation mode for the non-saturation typelinear amplification. In the first operation mode for the saturationtype nonlinear amplification, for example, the bias voltages Vgs1 andVgs2 are respectively set to such bias levels that large amplificationgains are obtained at the transistors Qn1 and Qn2 of the first-stageamplifier stage 10 and the final-stage amplifier stage 11. In the secondoperation mode for the non-saturation type linear amplification, thebias voltages Vgs1 and Vgs2 are respectively set to such bias levelsthat high operation linearity can be obtained at the transistors Qn1 andQn2 of the first-stage amplifier stage 10 and the final-stage amplifierstage 11.

In order to detect an RF transmit power level transmitted from theantenna, the directional coupler 13 is coupled via the output matchingcircuit (not shown) to the drain of the common-source N channel MOStransistor Qn2 of the final-stage amplifier stage 11. A main line of thedirectional coupler 13 is coupled between the output matching circuit(not shown) and the antenna. An RF signal proportional to the level ofthe RF transmit output signal P_(OUT) is generated from thecorresponding sub-line disposed close to the main line approximately inparallel. The directional coupler 13 has a high degree of couplingbetween the main line and the sub-line with respect to a traveling waveof the RF transmit output signal P_(OUT) from the output of the RF poweramplifier 100 to the antenna. However, the directional coupler 13 has arelatively low degree of coupling between the main line and the sub-linewith respect to a reflected wave of the RF transmit output signalP_(OUT) from the antenna to the output of the RF power amplifier 100 dueto a load mismatch at the antenna. Thus, the level of the RF signal fromthe sub-line of the directional coupler 13 is approximately determinedby the level of the traveling wave of the RF transmit output signalP_(OUT) from the output of the RF power amplifier 100 to the antenna.

Incidentally, the choke coils L1 and L2 used as the loads of thetransistors Qn1 and Qn2 of the first-stage amplifier stage 10 and thefinal-stage amplifier stage 11 are respectively comprised of an internallayer wiring of a multilayer wiring insulated substrate or board of apower module that forms the RF power amplifier 100. As anotherembodiment, however, these choke coils can also be configured byair-core coils located at the surface of the multilayer wiring insulatedsubstrate. Further, as an alternative thereto, they can also beconfigured by spiral coils located at the surface of the siliconsemiconductor chip.

The RF power amplifier 100 shown in FIG. 1 further includes a firstdetector 14, an adder 15, a low-pass filter 16, a second detector 17 andan adjustment circuit 18 formed in the silicon semiconductor chip.

The first detector 14 is used to adapt to a variation in load at theantenna of one RF power amplifier which executes the first operationmode for the saturation type nonlinear amplification and the secondoperation mode for the non-saturation type linear amplification. Thesecond detector 17 is used to reduce an adjacent channel leakage powerratio (ACPR) of a single type commonly-used RF power amplifier uponadaptation to a load variation in the second operation mode for thenon-saturation type linear amplification.

Since the RF signal formed from the sub-line of the directional coupler13 is supplied to the corresponding input terminal of the first detector14 for adapting to the load variation at the antenna, a first detectionsignal having a signal level of the traveling wave of the RF transmitoutput signal P_(OUT) sent from the output of the RF power amplifier 100to the antenna is formed from the corresponding output terminal of thefirst detector 14. The first detection signal is supplied to one inputterminal of the adder 15.

In an overload state at the antenna of the RF power amplifier, theoutput voltage at the drain of the common-source N channel MOStransistor Qn2 of the final-stage amplifier stage 11 increasessignificantly. In doing so, signal distortion of the MOS transistor Qn2in the second operation mode for the non-saturation type linearamplification also increases noticeably, thus causing the risk of theadjacent channel leakage power ratio (ACPR) being deterioratedsignificantly. Thus, the output voltage at the drain of thecommon-source N channel MOS transistor Qn2 of the final-stage amplifierstage 11 is supplied to its corresponding input terminal of the seconddetector 17 for reducing the adjacent channel leakage power ratio (ACPR)at the overload. Comparing with a relatively low first input thresholdvoltage Vth1 of the first detector 14, a second input threshold voltageVth2 of the second detector 17 is set to a relatively large voltagevalue. A second detection signal having an excessive output voltagelevel of the transistor Qn2 of the final-stage amplifier stage 11 in theoverload state at the antenna is formed from the corresponding outputterminal of the second detector 17. The second detection signal issupplied to the other input terminal of the adder 15.

The adder 15 performs analog addition of the analog voltages of both thefirst detection signal from the first detector 14, which is supplied tothe one input terminal, and the second detection signal from the seconddetector 17, which is supplied to the other input terminal via theadjustment circuit 18. A signal outputted from the adder 15 isattenuated in RF component and high-frequency noise by the low-passfilter 16 and supplied to the corresponding input terminal of the A/Dconverter 22 of the signal processing unit 200 as a gain control analogsignal. The A/D converter 22 converts the gain control analog signal toa gain control digital signal. The controller 23 sets the amplificationgain of the programmable gain amplifier 21 in response to the gaincontrol digital signal.

Thus, when the first detection signal outputted from the output terminalof the first detector 14, which is approximately proportional to thesignal level of the traveling wave of the RF transmit output signalP_(OUT) from the output of the RF power amplifier 100 to the antennabecomes high, the amplification gain of the programmable gain amplifier21 is reduced. In such a state that the antenna is brought to theoverload state and the excessive output voltage level is generated fromthe transistor Qn2 of the final-stage amplifier stage 11, theamplification gain of the programmable gain amplifier 21 is reduced inresponse to the second detection signal outputted from the outputterminal of the second detector 17.

<<Operation of RF Power Amplifier>> <<Adaptation to Load Variation atAntenna>>

A variation in load at the antenna occurs where the RF power amplifier100 shown in FIG. 1 executes the first operation mode for the saturationtype nonlinear amplification and the second operation mode for thenon-saturation type linear amplification. Since the antenna is normallyset to 50Ω, the output matching circuit between the drain of thecommon-source N channel MOS transistor Qn2 of the final-stage amplifierstage 11 and the antenna performs impedance matching between anextremely low output resistance of the drain of the transistor Qn2 andan impedance of 50Ω of the antenna.

Meanwhile, when the antenna of the cellular phone terminal approaches atable or the like such as a metal product, the impedance of the antennanormally set to 50Ω is reduced to a slightly low value. There are caseswhere a cellular phone terminal that supports a multimode communicationtransmits an RF transmit signal having a low band frequency ofapproximately 0.8 GHz to 1.0 GHz of GSM850 and GSM900 and transmits anRF transmit signal having a high band frequency of approximately 1.7 GHzto 2.0 GHz of DCS1800, PCS1900 and WCDMA 2100. Upon the transmission ofthe RF transmit signal having the low band frequency, the impedance ofthe drain of the transistor Qn2 of the final-stage amplifier stage 11 asthe output terminal of the RF power amplifier 100 is reduced to aslightly low value due to the approach of the antenna to the metal. Uponthe transmission of the RF transmit signal having the high bandfrequency, however, the impedance of the drain of the transistor Qn2 ofthe final-stage amplifier stage 11 as the output terminal of the RFpower amplifier 100 is increased to a slightly high value due to theimpedance of the output matching circuit with the approach of theantenna to the metal. As a result, the load impedance of the outputterminal of the RF power amplifier 100 fluctuates in a slight range dueto the load variation of the antenna.

With fluctuations in the slight range, in the load impedance of theoutput terminal of the RF power amplifier 100 that performs the firstoperation mode for the saturation type nonlinear amplification fortransmission of the GMSK constant envelope signal in the GSMcommunications, for example, the RF transmit output signal P_(OUT) fromthe output of the RF power amplifier 100 to the antenna also fluctuatesin a slight range. When, however, the first detection signal from theoutput terminal of the first detector 14, approximately proportional tothe signal level of the traveling wave of the RF transmit output signalP_(OUT) from the output of the RF power amplifier 100 to the antennabecomes high, the amplification gain of the programmable gain amplifier21 is reduced. As a result, even though the slight load variation at theantenna has occurred, the RF transmit output signal P_(OUT) suppliedfrom the output of the RF power amplifier 100 to the antenna can bemaintained approximately constant. As a result, characteristics similarto the suppression characteristics for the load variation by theparallel RF power amplifier descried in the non-patent document 6 can berealized by the RF power amplifier 100 shown in FIG. 1, which is of thesingle type and small in the number of elements.

<<Adaptation to Overload State at Antenna>>

On the other hand, when the antenna of the cellular phone terminal isbrought into contact with the table or the like such as the metalproduct, the impedance of the antenna normally set to 50Ω is reduced toa considerably low value. Incidentally, there are cases where a cellularphone terminal that supports a multimode communication transmits an RFtransmit signal having a low band frequency of approximately 0.8 GHz to1.0 GHz of GSM850 and GSM900 and transmits an RF transmit signal havinga high band frequency of approximately 1.7 GHz to 2.0 GHz of DCS1800,PCS1900 and WCDMA 2100. Upon the transmission of the RF transmit signalhaving the low band frequency, the impedance of the drain of thetransistor Qn2 of the final-stage amplifier stage 11 as the outputterminal of the RF power amplifier 100 is reduced to a considerably lowvalue near a short state due to the contact of the antenna with themetal. Upon the transmission of the RF transmit signal having the highband frequency, however, the impedance of the drain of the transistorQn2 of the final-stage amplifier stage 11 as the output terminal of theRF power amplifier 100 is increased to a considerably high value near anopen state due to the impedance of the output matching circuit with thecontact of the antenna to the metal. As a result, the load impedance ofthe output terminal of the RF power amplifier 100 changes in a widerange from about 17Ω to about 150Ω due to the load variation of theantenna.

FIG. 2 is a diagram for describing the operation of the RF poweramplifier 100 when the load impedance of the RF power amplifier 100shown in FIG. 1 changes. The horizontal and vertical axes in FIG. 2respectively indicate a drain-source voltage Vds and a drain current Idof the transistor Qn2 of the final-stage amplifier stage 11 in the RFpower amplifier 100 shown in FIG. 1. In FIG. 2, R, R+ΔR and R−ΔRrespectively indicate a state in which the input impedance (loadimpedance) of the output matching circuit as viewed from the drain ofthe transistor Qn2 of the final-stage amplifier stage 11 is a propervalue, a state in which the input impedance is a value larger than theproper value and a state in which the input impedance is a value smallerthan the proper value. Positive characteristics of the drain-sourcevoltage Vds/drain current Id of the transistor Qn2 dependent on thevalue of a gate-source voltage VgsQn2 of the transistor Qn2 of thefinal-stage amplifier stage 11 are also shown in FIG. 2. Incidentally,in FIG. 2, a drain current Id of a middle level flows in response to agate-source voltage VgsQn2 _(—M) of a middle level at the transistorQn2, a drain current Id of a high level flows in response to agate-source voltage VgsQn2 _(—H) of a high level at the transistor Qn2,and a drain current Id of a low level flows in response to a gate-sourcevoltage VgsQn2 _(—L) of a low level at the transistor Qn2.

It can be understood from FIG. 2 that when the load impedance of the RFpower amplifier 100 shown in FIG. 1 changes from the state R of beingthe proper value to the overload state R+ΔR, the amplification gaincorresponding to each of a change in the drain-source voltage Vds and achange in the drain current Id responsive to a change in the gate-sourcevoltage VgsQn2 is reduced in a region low in the drain-source voltageVds. When the RF power amplifier 100 shown in FIG. 1 is performing thesecond operation mode for the non-saturation type linear amplification,for example, the reduction in the amplification gain of the transistorQn2 in the region low in the drain-source voltage Vds of the transistorQn2 of the final-stage amplifier stage 11 causes an increase in signaldistortion and degradation in adjacent channel leakage power ratio(ACPR).

FIG. 3 is a diagram for describing the dependence of a drain-sourcevoltage Vds of the transistor Qn2 of the final-stage amplifier stage 11where the load impedance of the RF power amplifier 100 shown in FIG. 1is brought a state R−ΔR of being a value smaller than a proper value, astate R of being the proper value and a state R+ΔR of being a valuelarger than the proper value. The horizontal and vertical axes shown inFIG. 3 respectively indicate the RF transmit input signal P_(IN) of theRF power amplifier 100 shown in FIG. 1 and the drain-source voltage Vdsof the transistor Qn2 of the final-stage amplifier stage 11. As shown inFIG. 3, the drain-source voltage Vds of the transistor Qn2 of thefinal-stage amplifier stage 11 increases approximately linearly when thelevel of the RF transmit input signal P_(IN) increases. When, however,the level of the RF transmit input signal P_(IN) becomes excessive, theincrease in the drain-source voltage Vds is gradually saturated andthereafter completely saturated. The saturation first starts in thestate R+ΔR in which the load impedance of the RF power amplifier 100 isof the value larger than the proper value. Next, the saturation startsin the state R in which the load impedance is of the proper value.Finally, the saturation starts in the state R−ΔR in which the loadimpedance is of the value smaller than the proper value.

FIG. 4 is a diagram for describing the dependence of an RF transmitoutput signal P_(OUT) of the RF power amplifier 100 where the loadimpedance of the RF power amplifier 100 shown in FIG. 1 is brought to astate R−ΔR of being a value smaller than a proper value, a state R ofbeing the proper value and a state R+ΔR of being a value larger than theproper value.

The RF transmit output signal P_(OUT) corresponds to output power.Assuming that the output impedance of the transistor Qn2 of thefinal-stage amplifier stage 11 is R_(OUT), the load impedance of the RFpower amplifier 100 is R and the drain current of the transistor Qn2 isId, the RF transmit output signal P_(OUT) is given by the followingequation:

$\begin{matrix}{P_{OUT} = {\frac{R_{OUT}}{R + R_{OUT}}*i_{d}*R^{2}}} & (1)\end{matrix}$

As well known, when the value of the output impedance R_(OUT) and thevalue of the load impedance R are equal to each other, the maximum valueof the RF transmit output signal P_(OUT) is obtained, whereas when theyare different from each other, the RF transmit output signal P_(OUT) islowered from the maximum value. Thus, as shown in FIG. 4, the RFtransmit output signal P_(OUT) of the non-saturated state in the state R(state in which it is equal to the value of the output impedanceR_(OUT)) in which the load impedance of the RF power amplifier 100 is ofthe proper value, becomes maximum. As shown in FIG. 4, even in the stateR−ΔR in which the load impedance of the RF power amplifier 100 is of thevalue smaller than the proper value and the state R+ΔR in which the loadimpedance is of the value larger than the proper value, the RF transmitoutput signal P_(OUT) in the non-saturated state is reduced.

Assuming that a change in the load impedance of the RF power amplifier100 due to the load variation is ΔR, a variation in the RF transmitoutput signal P_(OUT) OUT corresponding to the output power is given bythe following equation:

$\begin{matrix}{{\Delta\; P_{OUT}} = {20\;{\log\left\lbrack {\frac{R_{OUT}}{R + {\Delta\; R} + R_{OUT}}\left( \frac{R + {\Delta\; R}}{R} \right)^{2}} \right\rbrack}}} & (2)\end{matrix}$

It has been demonstrated by discussions of the present inventors thatthe adjacent channel leakage power ratio (ACPR) corresponding to thesignal distortion increases abruptly from immediately before the RFtransmit output signal P_(OUT) corresponding to the output power shownin FIG. 4 starts to saturate.

FIG. 5 is a diagram for describing the dependence of the start ofsaturation of the RF transmit output signal P_(OUT) of the RF poweramplifier 100 and an abrupt increase in adjacent channel leakage powerratio (ACPR) where the load impedance of the RF power amplifier of FIG.1 is brought to a state R−ΔR of being a value smaller than a propervalue, a state R of being the proper value, and a state R+ΔR of being avalue larger than the proper value.

In FIG. 5, a thick broken line R indicates a saturation characteristicof an RF transmit output signal P_(OUT) with respect to an RF transmitinput signal P_(IN) in the state R in which the load impedance is of theproper value. A thick solid line ACPR@R indicates a characteristic ofdependence of an adjacent channel leakage power ratio (ACPR) on an RFtransmit input signal P_(IN) in the state R in which the load impedanceis of the proper value. In the state R in which the load impedance is ofthe proper value, the RF transmit output signal P_(OUT) OUT starts tosaturate from an RF transmit input signal P_(IN) of a middle level andthe adjacent channel leakage power ratio (ACPR) starts to increaseabruptly.

In FIG. 5, a thin broken line R−ΔR indicates a saturation characteristicof an RF transmit output signal P_(OUT) with respect to an RF transmitinput signal P_(IN) in the state R−ΔR in which the load impedance is ofthe value smaller than the proper value. A thin solid line ACPR@R−ΔRindicates a characteristic of dependence of an adjacent channel leakagepower ratio (ACPR) on an RF transmit input signal P_(IN) in the stateR−ΔR in which the load impedance is of the value smaller than the propervalue. In the state R−ΔR in which the load impedance is of the valuesmaller than the proper value, the RF transmit output signal P_(OUT)starts to saturate beginning with an RF transmit input signal P_(IN) ofa high level, and the adjacent channel leakage power ratio (ACPR) startsto increase abruptly.

In FIG. 5, a thin broken line R+ΔR indicates a characteristic ofsaturation of an RF transmit output signal P_(OUT) with respect to an RFtransmit input signal P_(IN) in the state R+ΔR in which the loadimpedance is of the value larger than the proper value. A thin sold lineACPR@R+ΔR indicates a characteristic of dependence of an adjacentchannel leakage power ratio (ACPR) on an RF transmit input signal P_(IN)in the state R+ΔR in which the load impedance is of the value largerthan the proper value. In the state R+ΔR in which the load impedance isof the value larger than the proper value, the RF transmit output signalP_(OUT) starts to saturate easier and the adjacent channel leakage powerratio (ACPR) starts to increase abruptly. As a result, when the loadimpedance of the RF power amplifier shown in FIG. 1 assumes highimpedance indicative of an overload state and close to an open state,the RF transmit output signal P_(OUT) starts to saturate easily even atan RF transmit input signal P_(IN) of a low level, and the adjacentchannel leakage power ratio (ACPR) starts to increase suddenly. It hasbeen demonstrated by discussions of the present inventors that thecharacteristic of the abrupt increase in ACPR in the overload state isextremely harmful to the second operation mode for the non-saturationlinear amplification for the transmission of the envelop change signalof 3π/8-8PSK in the EDGE communications or HPSK in the WCDMAcommunications. With the abrupt increase in ACPR in the overload state,signal distortion of the envelope change signal in the EDGE or WCDMAcommunications increases significantly.

On the other hand, it is understood from FIG. 3 that in the overloadstate indicative of the state R+ΔR in which the load impedance is of thevalue larger than the proper value, the drain-source voltage Vds of thetransistor Qn2 of the final-stage amplifier stage 11 increasessignificantly as compared with the drain-source voltage in the state inwhich the load impedance is of the proper value.

In the RF power amplifier 100 shown in FIG. 1, however, the outputvoltage of the drain of the common-source N channel MOS transistor Qn2of the final-stage amplifier stage 11 has been supplied to the inputterminal of the second detector 17 having the second input thresholdvoltage Vth2 set to the relatively large voltage value. Thus, in the RFpower amplifier 100 shown in FIG. 1, a high-level overload detectionsignal is detected from the output terminal of the second detector 17 inthe overload state indicative of the state R+ΔR in which the loadimpedance is of the value larger than the proper value. The high-leveloverload detection signal detected from the output terminal of thesecond detector 17 is supplied to the input terminal of the A/Dconverter 22 of the signal processing unit 200 via the adjustmentcircuit 18, adder 15 and low-pass filter 16. The A/D converter 22converts an overload detection analog signal to an overload detectiondigital signal. The controller 23 reduces the amplification gain of theprogrammable gain amplifier 21 in response to the overload detectiondigital signal. Incidentally, the adjustment circuit 18 adjusts thelevel of the overload detection signal outputted from the seconddetector 17 and sets a coefficient a to a constant of 1 or less.

Thus, the level of the RF transmit input signal P_(IN) supplied to theinput terminal of the RF power amplifier shown in FIG. 1 is reduced withthe reduction in the amplification gain of the programmable gainamplifier 21. The reduction in the amplification gain of theprogrammable gain amplifier 21 and the reduction in the signal of the RFtransmit input signal P_(IN) are executed until the high-level overloaddetection signal is not formed from the output terminal of the seconddetector 17. Thus, even in the overload state indicative of the stateR+ΔR in which the load impedance is of the value larger than the propervalue, an RF transmit input signal P_(IN) of a signal level sufficientlylower that at the saturation start point of the RF transmit outputsignal P_(OUT) shown in FIG. 4 is supplied to the input terminal of theRF power amplifier 100. In doing so, the RF power amplifier 100 shown inFIG. 1 is capable of reducing the abrupt increase in the adjacentchannel leakage power ratio (ACPR) with the start of saturation of theRF transmit output signal P_(OUT). As a result, it is possible to reducethe abrupt increases in the adjacent channel leakage power ratio (ACPR)and the signal distortion in the second operation mode for thenon-saturation type linear amplification for the transmission of theenvelope change signal of 3π/8-8PSK in the EDGE communications or HPSKin the WCDMA communications in the overload state.

<<First Detector and Second Detector>>

FIG. 6 is a diagram showing input/output characteristics of the firstand second detectors 14 and 17 of the RF power amplifier 100 of FIG. 1.

The horizontal and vertical axes shown in FIG. 6 respectively indicatethe input and output voltages of the first and second detectors 14 and17. As described above, the second input threshold voltage Vth2 of thesecond detector 17 has been set to the relatively large voltage valuewhen compared with the relatively low first input threshold voltage Vth1of the first detector 14. Thus, the level of the output voltage(14(V_(DET1))) of the first detection signal from the first detector 14becomes higher than that of the output voltage (17(V_(DET2))) of thesecond detection signal from the second detector 17 with respect to thecorresponding input voltage In of the same level. The output voltage(18(aV_(DET2))) of the adjustment circuit 18 having the coefficient aset to the constant of 1 or less is also shown in FIG. 6.

FIG. 7 is a diagram showing the configurations of the first and seconddetectors 14 and 17 of the RF power amplifier 100 shown in FIG. 1.

As shown in FIG. 7, the first detector 14 comprises a multistageamplifier circuit including a first stage amplifier circuit 141, asecond stage amplifier circuit 142 and a third stage amplifier circuit143, a multistage detection circuit including a first stage detectioncircuit 144, a second stage detection circuit 145 and a third stagedetection circuit 146, and an addition circuit 147. The first detector14 can have high detection sensitivity owing to the multistage amplifiercircuit and the multistage detection circuit of the first detector 14.

As shown in FIG. 7, the second detector 17 comprises a single stageamplifier circuit 171 and a single stage detection circuit 172 and canhave relatively low detection sensitivity. An output level of the seconddetector 17 can be adjusted by the coefficient a set to the adjustmentcircuit 18. Further, the relatively large second input threshold voltageVth2 set to the second detector 17 can be adjusted by at least one of athreshold voltage of an amplifier transistor of the single stageamplifier circuit 171 and a threshold voltage of a detection diode ofthe single stage detection circuit 172.

<<Output Matching Circuit and Directional Coupler>>

FIG. 8 is a diagram showing one example illustrative of an outputmatching circuit OMN and a directional coupler 13′ coupled to the drainoutput electrode of the common-source N channel MOS transistor Qn2 ofthe final-stage amplifier stage 11 in the RF power amplifier 100 shownin FIG. 1.

The output matching circuit OMN comprises strip lines SL1 and SL2 thatconfigure inductors, and capacitors C3, C4 and C5. An input terminal ofthe output matching circuit OMN is coupled to the drain output electrodeof the MOS transistor Qn2 of the final-stage amplifier stage 11. An RFtransmit output signal P_(OUT) outputted from an output terminal of theoutput matching circuit OMN is supplied to the antenna of the cellularphone terminal. The strip line SL1 on the input terminal side of theoutput matching circuit OMN also functions as a main line of thedirectional coupler 13′.

The directional coupler 13′ comprises a strip line SL3 that forms asub-line thereof and a terminal resistor Z. The main line SL1 andsub-line SL3 of the directional coupler 13′ are disposed close to eachother approximately in parallel. One end of the sub-line SL3 is suppliedwith an RF amplifier signal of the drain output electrode of the MOStransistor Qn2 through a capacitor C2. The other end of the sub-line SL3is coupled to a ground voltage through the terminal resistor Z. An RFdetection signal generated from one end of the sub-line SL3 is suppliedto the input terminal of the first detector 14. The RF detection signalgenerated from one end of the sub-line SL3 is proportional to the signallevel of a traveling wave of the RF transmit output signal P_(OUT).

<<Adjustment Circuit having Nonlinear Characteristic>>

FIG. 9 is a diagram showing a configuration of an RF power amplifier inwhich the adjustment circuit 18 mounted to the RF power amplifier 100shown in FIG. 1 and having the coefficient a set to the constant of 1 orless is replaced with an adjustment circuit 18′ having a nonlinearcharacteristic. The RF power amplifier shown in FIG. 9 is similar inother configuration to the RF power amplifier shown in FIG. 1.

The adjustment circuit 18′ mounted to the RF power amplifier 100 shownin FIG. 9 has a nonlinear transfer function f(V_(DET2)) defined by asecondary function or a function of higher order than it or anexponential function. Thus, as the level of an output voltage(17(V_(DET2))) of a second detection signal from a second detector 17increases, an output voltage (18′f(V_(DET2))) having a level higher thanthat of the output voltage (18(aV_(DET2))) of the adjustment circuit 18shown in FIG. 1 can be generated from the adjustment circuit 18′ havingthe nonlinear characteristic.

Thus, as the level of the output voltage (17(V_(DET2))) of the seconddetection signal outputted from the second detector 17 in an overloadstate becomes higher than the second input threshold voltage Vth2, thelevel of the output voltage (18′f(V_(DET2))) outputted from theadjustment circuit 18′ further increases. As a result, abrupt increasesin adjacent channel leakage power ratio (ACPR) and signal distortion inthe second operation mode for the non-saturation type linearamplification for the transmission of the envelope change signal in theEDGE or WCDMA communications in the overload state can be reducedfurther effectively.

<<Adaptation to Overcurrent State>>

In order to adapt to the antenna load variation as described above, theamplification gain of the programmable gain amplifier 21 is controlledin accordance with the signal level of the traveling wave of the RFtransmit output signal P_(OUT) from the RF power amplifier 100 in thefirst operation mode for the saturation type nonlinear amplification forthe transmission of the constant envelope signal in the GSMcommunications to thereby maintain the RF transmit output signal P_(OUT)approximately constant.

As compared with the WCDMA and EDGE communications, however, the maximumRF transmit output signal P_(OUT(MAX)) in the GSM communications, of theRF power amplifier that performs the first operation mode for thesaturation type nonlinear amplification to transmit the constantenvelope signal is extremely large like an approximately 2.0 watts inthe first operation mode for the saturation type nonlinearamplification.

Meanwhile, the impedance of the drain of the transistor Qn2 of thefinal-stage amplifier stage 11 in the RF power amplifier 100 of FIG. 1is reduced to a considerably low value near a short state due to thecontact of the antenna with the metal upon transmission of the RFtransmit signal extremely large in constant envelope signal in the GSMcommunications at the low band frequency. In doing so, the drain currentId of the transistor Qn2 of the final-stage amplifier stage 11 isbrought to an extremely large overcurrent state. Thus, the wearing outof a battery of the cellular phone terminal becomes noticeable and therisk of the transistor Qn2 of the final-stage amplifier stage 11 beingdestroyed also occurs. Accordingly, it has been demonstrated bydiscussions of the present inventors that it is necessary to adapt tothe overcurrent state of the output transistor of the final-stageamplifier stage due to the load short-circuit of the RF power amplifierthat performs the saturation type nonlinear amplification to transmitthe constant envelope signal in the GSM communications.

FIG. 10 is a diagram showing a configuration of an RF power amplifierwherein an overcurrent protection circuit for adapting to a loadshort-circuit at the transmission of a large RF transmit signal in theGSM communications is added to the RF power amplifier 100 shown inFIG. 1. The RF power amplifier shown in FIG. 10 is similar in otherconfiguration to the RF power amplifier shown in FIG. 1.

The overcurrent protection circuit added to the RF power amplifier 100shown in FIG. 10 includes a current sense circuit 19, an overcurrentdetection circuit 18B and an adder 15A. The current sense circuit 19comprises a current sense N channel MOS transistor Qn3 whose gatecontrol input terminal is supplied with a gate control input signal of acommon-source N channel MOS transistor Qn2 of a final-stage amplifierstage 11 of the RF power amplifier 100, and a current sense resistorRcs. The source of the current sense transistor Qn3 is coupled to aground voltage and the drain thereof is coupled to a source voltage Vddthrough the current sense resistor Rcs. An element size of the currentsense transistor Qn3 is set considerably smaller than an element size ofthe RF amplifier transistor Qn2, whereby a sense current flowing throughthe current sense transistor Qn3 at the load short-circuit can bereduced.

Even at the normal operation and the load short-circuit, the sensecurrent flowing through the transistor Qn3 is converted to a sensevoltage by the current sense resistor Rcs. The sense voltage appliedbetween both ends of the current sense resistor Rcs is monitored by theovercurrent detection circuit 18B. For example, the overcurrentdetection circuit 18B has an input threshold voltage. When the sensevoltage of the current sense resistor Rcs exceeds the input thresholdvoltage, the level of an overcurrent detection output signal becomeshigh. The high-level overcurrent detection output signal outputted fromthe overcurrent detection circuit 18B is transmitted to thecorresponding A/D converter 22 of the signal processing unit 200 via theadder 15A, adder 15B and low-pass filter 16. The controller 23 reducesthe amplification gain of the programmable gain amplifier 21 in responseto an overcurrent detection digital signal outputted from the A/Dconverter 22.

Thus, the increases in the sense current flowing through the currentsense transistor Qn3 at the load short-circuit and RF current flowingthrough the RF amplifier transistor Qn2 of the final-stage amplifierstage 11 are stopped in such a manner that the sense voltage of thecurrent sense resistor Rcs does not exceed the input threshold voltageof the overcurrent detection circuit 18B. As a result, the overcurrentprotection circuit including the current sense circuit 19, theovercurrent detection circuit 18B and the adder 15A performs anovercurrent protecting operation of current limiter type in the RF poweramplifier 100 shown in FIG. 10.

As another embodiment, when the sense voltage of the current senseresistor Rcs exceeds the input threshold voltage of the overcurrentdetection circuit 18B once upon the load short-circuit, a latch circuitcan be set from a reset state indicative of an initial state to a setstate based on overcurrent detection by the high-level overcurrentdetection output signal outputted from the overcurrent detection circuit18B. The controller 23 is controlled by a signal outputted from thelatch circuit set to the set state to thereby make it possible to reducethe amplification gain of the programmable gain amplifier 21 to zero.Such an overcurrent protection circuit performs an overcurrentprotecting operation of shutdown type. Its subsequent operationresumption is enabled by changing the latch circuit from the set stateto the reset state.

<<Independent Feedback>>

FIG. 11 is a diagram showing a configuration of an RF power amplifierwherein in the RF power amplifier 100 shown in FIG. 1, first feedbackfrom the output of the first detector 14 to the controller 23 of thesignal processing unit 200 and second feedback from the output of thesecond detector 17 to the controller 23 of the signal processing unit200 are rendered independent of each other. The RF power amplifier shownin FIG. 11 is similar in other configuration to the RF power amplifiershown in FIG. 1.

The first feedback comprised of a low-pass filter 16B and an A/Dconverter 22B is used to stabilize an RF transmit output signal P_(OUT)adaptable to a load variation The second feedback comprised of anadjustment circuit 18, a low-pass filter 16A and an A/D converter 22A isused to reduce abrupt increases in the adjacent channel leakage powerratio (ACPR) and signal distortion in the second operation mode for thenon-saturation type linear amplification upon transmission of theenvelope change signal in the EDGE or WCDMA communications in theoverload state.

Since the first feedback and the second feedback are independentlydisposed in the RF power amplifier 100 shown in FIG. 11, the present RFpower amplifier is simple as compared with the RF power amplifier inwhich the two feedback have been combined as shown in FIG. 1, anadvantage is brought about in that circuit design becomes easy.

<<Configuration of Cellular Phone>>

FIG. 12 is a diagram showing a configuration of a cellular phone inwhich an RF power amplifier 100 according to an embodiment of thepresent invention and a signal processing unit 200 are provided as adual-band configuration that covers low and high band frequencies, andwhich includes a DC/DC converter 300, an antenna switch 400, duplexersor the like and an antenna ANT. In the cellular phone having theconfiguration shown in FIG. 12, any of the RF power amplifier shown inFIG. 1, the RF power amplifier shown in FIG. 9, the RF power amplifiershown in FIG. 10 and the RF power amplifier shown in FIG. 11 can beadopted as the RF power amplifier 100.

The DC/DC converter 300 is of a back-boost converter and can cover aback operation mode and a boost operation mode. With the supply of abattery voltage Vbat from a battery mounted to the cellular phone to theDC/DC converter 300, a source or power supply voltage Vdd supplied tothe RF power amplifier 100 is generated from the DC/DC converter 300.When a high RF transmit output signal P_(OUT) in the GSM communicationsis transmitted when the battery voltage Vbat from the battery is low,the DC/DC converter 300 is set to the boost operation mode. When a lowRF transmit output signal P_(OUT) is transmitted where the batteryvoltage Vbat from the battery is high in reverse, the DC/DC converter300 is set to the back operation mode, thereby making it possible toimprove power added efficiency (PAE). Incidentally, PAE is anabbreviation of Power Added Efficiency.

The antenna switch 400 comprises a single pole 7 throw (SP7T) typeswitch. In the field of the antenna switch, a common input/outputterminal I/O coupled to the antenna ANT is called “Single Pole”, andother terminals such as transmission terminals, reception terminals,transmission/reception terminals are called “Throw”.

Incidentally, the low band frequency is of an RF transmit signalfrequency ranging from approximately 0.8 GHz to 1.0 GHz of GSM850 andGSM900. The high band frequency is of an RF transmit signal frequencycorresponding to a high band frequency ranging from approximately 1.7GHz to 2.0 GHz of DCS1800, PCS1900 and WCDMA2100.

The RF power amplifier 100 shown in FIG. 12 includes a first-stageamplifier stage 10_LB, a final-stage amplifier stage 11_LB, adirectional coupler 13_LB, a first detector 14_LB, an adder 15_LB, alow-pass filter 16_LB, and a second detection/adjustment circuit 17,18_LB, which cover the low band frequency. These circuits that cover thelow band frequency in the RF power amplifier shown in FIG. 12 areoperated in a manner similar to the RF power amplifier 100 shown in FIG.1 as to the stabilization of an RF transmit output signal P_(OUT) OUTadaptable to a load variation and a reduction in the increase of anadjacent channel leakage power ratio in a second operation mode at thetransmission of an envelope change signal in an overload state. As aresult, a lowband gain control signal PGAcnt_LB for controlling theamplification gain of a lowband programmable gain amplifier that coversthe low band frequency of the signal processing unit 200 is generatedfrom the circuits that cover the low band frequency of the RF poweramplifier 100 shown in FIG. 12.

The RF power amplifier 100 shown in FIG. 12 includes a first-stageamplifier stage 10_HB, a final-stage amplifier stage 11_HB, adirectional coupler 13_HB, a first detector 14_HB, an adder 15_HB, alow-pass filter 16_HB and a second detection/adjustment circuit 17,18_HB, which cover the high band frequency. These circuits that coverthe high band frequency in the RF power amplifier 100 shown in FIG. 12are operated in a manner similar to the RF power amplifier 100 shown inFIG. 1 as to the stabilization of the RF transmit output signal P_(OUT)OUT adaptable to the load variation and the reduction in the increase ofthe adjacent channel leakage power ratio in the second operation mode atthe transmission of the envelope change signal in the overload state. Asa result, a highband gain control signal PGAcnt_HB for controlling theamplification gain of a highband programmable gain amplifier that coversthe high band frequency of the signal processing unit 200 is generatedfrom the circuits that cover the high band frequency of the RF poweramplifier 100 shown in FIG. 12.

The signal processing unit 200 shown in FIG. 12 generates an RF transmitsignal of a low band frequency to the RF power amplifier 100 from itslower part. As the RF transmit signal of the low band frequency, theremay be mentioned the following: RF transmit signals TX of GSM850 andGSM900: GSM850 and GSM900, and RF transmit signal TX of WCDMA at BAND 5:BAND5. The frequency of the RF transmit signal of GSM850 is set to 824to 849 MHz. The frequency of the RF transmit signal of GSM900 is set to889 to 915 MHz. The frequency of the RF transmit signal of WCDMA atBAND5 is set to 824 to 849 MHz exactly identical to GSM850. The RFtransmit signals of GSM850 and GSM900 and the RF transmit signal ofWCDMA at BAND5 are selected by a signal pole 2 throw (SP2T) type switchSW2 and supplied to the input terminal of the first-stage amplifierstage 10_LB that covers the low band frequency of the RF power amplifier100 shown in FIG. 12. A lowband RF amplifier output signal sent from theoutput terminal of the final-stage amplifier stage 11_LB that covers thelow band frequency of the RF power amplifier 100 shown in FIG. 12 issupplied to a single pole 2 throw type switch SW4 via the directionalcoupler 13_LB. The lowband RF amplifier output signal for each of GSM850and GSM900 is selected by the switch SW4 and supplied to the antenna ANTvia the single pole 7 throw type antenna switch 400. The lowband RFamplifier output signal for the WCDMA of BAND5 is selected by the switchSW4 and supplied to the antenna ANT via the duplexer Duplx_B5 and theantenna switch 400.

The signal processing unit 200 shown in FIG. 12 generates an RF transmitsignal of a high band frequency to the RF power amplifier 100 from itscentral part. As the RF transmit signal of the high band frequency,there may be mentioned the following: RF transmit signals TX for DCS1800and PCS1900 in GSM system: DCS1800 and PCS1900, and RF transmit signalsTX for BAND1 and BAND2 in WCDMA system: BAND1 and BAND2. The frequencyof the RF transmit signal of DCS1800 in the GSM system is set to 1710 to1785 MHz, and the frequency of the RF transmit signal of PCS1900 in theGSM system is set to 1850 to 1910 MHz. The frequencies of the RFtransmit signals corresponding to RF transmit signals TX in the WCDMAsystem: BAND1 and BAND2 are respectively set to 1920 to 1980 MHz and1850 to 1910 MHz. The RF transmit signals of DCS1800 and PCS1900 in theGSM system and the RF transmit signals TX corresponding to the RFtransmit signals in the WCDMA system: BAND1 and BAND2 are selected by asingle pole 3 throw type switch SW1 and supplied to the input terminalof the first-stage amplifier stage 10_HB that covers the high bandfrequency of the RF power amplifier 100 shown in FIG. 12. A lowband RFamplifier output signal from the output terminal of the final-stageamplifier stage 11_HB that covers the high hand frequency of the RFpower amplifier 100 shown in FIG. 12 is supplied to a single pole 3throw type switch SW3 via the directional coupler 13_HB. A highband RFamplifier output signal for each of DCS1800 and PCS1900 in the GSMsystem is selected by the switch SW3 and supplied to the antenna ANTthrough the single pole 7 throw type antenna switch 400. The highband RFamplifier output signals for BAND1 and BAND2 of WCDMA are selected bythe switch SW3 and supplied to the antenna ANT via the duplexerDuplx_B1, duplexer Duplx_B2 and antenna switch 400.

The signal processing unit 200 shown in FIG. 12 is supplied with RFreceive signals of low band and high band frequencies received by theantenna ANT from its upper part. As the RF receive signals for the lowband frequency, there may be mentioned the following: RF receive signalsRX of GSM850 and GSM900: GSM850 and GSM900 and RF receive signal RX ofWCDMA at BAND5: BAND5. The frequency of the RF receive signal of GSM850is set to 869 to 894 MHz, and the frequency of the RF receive signal ofGSM900 is set to 925 to 950 MHz. The frequency of the RF receive signalof WCDMA at BAND5 are set to 869 to 894 MHz exactly identical to GSM850.The RF receive signals of GSM850 and GSM900 are supplied to the signalprocessing unit 200 via the single pole 7 throw type antenna switch 400and a surface acoustic wave filer SAW1. The RF receive signal of WCDMAat BAND5 is supplied to the signal processing unit 200 via the singlepole 7 throw antenna switch 400 and duplexer Duplx_B5.

The RF receive signals for the high band frequency are as follows: RFreceive signals RX of DCS1800 and PCS1900 in GSM system: DCS1800 andPCS1900, and RF receive signals RX in WCDMA system at BAND1 and BAND2:BAND1 and BAND2. The frequency of the RF receive signal of DCS1800 inthe GSM system is set to 1805 to 1850 MHz. The frequency of the RFreceive signal of PCS1900 in the GSM system is set to 1930 to 1990 MHz.The frequencies of the RF receive signals corresponding to the RFreceive signals RX in the WCDMA system: BAND1 and BAND2 are respectivelyset to 2110 to 2170 MHz and 1930 to 1990 MHz. The RF receive signals ofDCS1800 and PCS1900 are supplied to the signal processing unit 200 viathe single pole 7 throw antenna switch 400 and a surface acoustic wavefilter SAW2. The RF receive signals of WCDMA at BAND1 and BAND2 aresupplied to the signal processing unit 200 via the single pole 7 throwtype antenna switch 400 and the duplexers Duplx_B1 and Duplx_B2.

While the invention made above by the present inventors has beendescribed specifically on the basis of the preferred embodiments, thepresent invention is not limited to the embodiments referred to above.It is needless to say that various changes can be made thereto withoutthe scope not departing from the gist thereof.

For example, the directional coupler for detecting the level of the RFtransmit output signal P_(OUT) of the RF power amplifier can be replacedwith a simple capacitor. An AC sense signal based on the capacitor isdetected by the envelope detector as described in the non-patentdocument 4. A detected voltage is compared with a reference voltage bythe error amplifier, and a bias current of a drive stage for driving afinal stage by the output of the error amplifier can also be controlled.

The RF power amplifier 100 can be comprised of a multistage amplifierstage having three or more stages, in which an intermediate-stageamplifier stage is coupled between the first-stage amplifier stage 10and the final-stage amplifier stage 11.

In the above embodiment of the present invention, the amplification gainof the programmable gain amplifier of the signal processing unit 200 hasbeen controlled upon stabilization of the RF transmit output signalP_(OUT) OUT adaptable to the load variation and the reduction in theincrease in adjacent channel leakage power ratio in the second operationmode at the transmission of the envelop change signal in the overloadstate. The present invention is not limited to this system. Upon such acase, the control bias voltage at the gate control electrode or basecontrol electrode of at least either one of the transistors of thefirst-stage amplifier stage 10 and the final-stage amplifier stage 11can also be controlled to control the amplification gain of the RF poweramplifier. That is, when the RF transmit output signal P_(OUT) increasesdue to the load variation and the adjacent channel leakage power ratioincreases in the second operation mode at the transmission of theenvelope change signal in the overload state, the control bias voltageof the transistor is reduced and the amplification gain of the RF poweramplifier is reduced.

Further, it is recommended that when each of the common-source N channelMOS transistors of the first-stage amplifier stage 10 and thefinal-stage amplifier stage 11 is replaced with a common-emitter NPNtype HBT, a base ballast resistor or an emitter ballast resistor iscoupled to the HBT to prevent thermal runaway of the HBT.

1. An RF power amplifier comprising: a final-stage amplifier stage whichamplifies an RF signal to generate an RF transmit output signal suppliedto an antenna of a communication terminal; and a signal detector whichdetects a level of the RF transmit output signal, wherein thefinal-stage amplifier stage includes a final stage transistor forgenerating the RF transmit output signal at an output electrode thereof,and a final-stage load element coupled between the output electrode ofthe final stage transistor and a source voltage, and executes a firstoperation mode for saturation type nonlinear amplification and a secondoperation mode for non-saturation type linear amplification, wherein aninput terminal of the signal detector is coupled to the output electrodeof the final stage transistor to thereby generate an RF detection signalresponsive to the level of the RF transmit output signal from an outputterminal of the signal detector, the RF power amplifier furtherincluding a first detector, a second detector and a control circuit,wherein an input terminal of the first detector is supplied with the RFdetection signal generated from the output terminal of the signaldetector to thereby generate a first detection signal from an outputterminal of the first detector, wherein an input terminal of the seconddetector is coupled to the output electrode of the final stagetransistor to thereby generate a second detection signal from an outputterminal of the second detector, wherein the first detection signalgenerated from the output terminal of the first detector contains afirst detection component responsive to a variation in the level of theRF transmit output signal due to a variation in load at the antenna whenthe RF power amplifier operates in the first operation mode for thesaturation type nonlinear amplification, wherein the second detectionsignal generated from the output terminal of the second detectorcontains a second detection component responsive to an increase inoutput voltage of the output electrode of the final stage transistor dueto an overload state of the antenna when the RF power amplifier operatesin the second operation mode for the non-saturation type linearamplification, wherein the first detector and the second detector have afirst input threshold voltage and a second input threshold voltagerespectively, wherein a level of the second input threshold voltage isset higher than a level of the first input threshold voltage, whereinthe first detection signal generated from the output terminal of thefirst detector and the second detection signal generated from the outputterminal of the second detector are supplied to the control circuit,wherein the control circuit controls the levels of the RF transmitoutput signal and the output voltage at the output electrode of thefinal stage transistor, wherein first feedback control by the firstdetector and the control circuit reduces the variation in the RFtransmit output signal generated from the output electrode of the finalstage transistor responsive to the load variation of the antenna uponthe operation in the first operation mode, and wherein second feedbackcontrol by the second detector and the control circuit reduces theincrease in the output voltage of the output electrode of the finalstage transistor responsive to the overload state of the antenna uponthe operation in the second operation mode.
 2. The RF power amplifieraccording to claim 1, wherein in the first operation mode for thesaturation type nonlinear amplification, the RF signal is of a constantenvelope signal in GSM communications, and wherein in the secondoperation mode for the non-saturation type linear amplification, the RFsignal is of an envelope change signal in either one of EDGE and WCDMAcommunications.
 3. The RF power amplifier according to claim 2, whereinthe signal detector is a directional coupler having a main line and asub-line disposed close to each other approximately in parallel.
 4. TheRF power amplifier according to claim 3, wherein an output matchingcircuit is coupled to the output electrode of the final stagetransistor, wherein the main line of the directional coupler is coupledbetween the output matching circuit and the antenna, and wherein an RFdetection signal response to the level of the RF transmit output signalis generated from the sub-line of the directional coupler.
 5. The RFpower amplifier according to claim 4, further including a first-stageamplifier stage, wherein the RF signal amplified by the final-stageamplifier stage is generated from the first-stage amplifier stage,wherein the first-stage amplifier stage includes a first stagetransistor which generates the RF signal, wherein a programmable gainamplifier for supplying an RF transmit input signal is coupled to acontrol input terminal of the first stage transistor, and wherein thecontrol circuit controls programmable gain of the programmable gainamplifier to thereby control the levels of the RF transmit output signaland the output voltage at the output electrode of the final stagetransistor.
 6. The RF power amplifier according to claim 3, wherein thecontrol circuit includes an adder which adds a component of the firstdetection signal generated from the output terminal of the firstdetector and a component of the second detection signal generated fromthe output terminal of the second detector, and a low-pass filtercoupled to the output of the adder, and wherein an analog control signalof the control circuit which controls the levels of the RF transmitoutput signal and the output voltage at the output electrode of thefinal stage transistor is generated from the output of the low-passfilter.
 7. The RF power amplifier according to claim 6, wherein thefirst stage transistor of the first-stage amplifier stage and the finalstage transistor of the final-stage amplifier stage are respectivelyeither one of an LDMOS transistor and a hetero bipolar transistor. 8.The RF power amplifier according to claim 7, further including a currentsense circuit and an overcurrent detection circuit, wherein the currentsense circuit includes a current sense transistor which has an elementsize smaller than the final stage transistor of the final-stageamplifier stage and which allows a sense current smaller than a currentflowing through the final stage transistor to flow therethrough, whereinthe overcurrent detection circuit supplies an overcurrent detectionsignal to the control circuit in response to the sense current flowingthrough the current sensor transistor of the current sense circuit, andwherein when the current flowing through the final stage transistor isbrought to an overcurrent state due to the load variation of the antennawhen the RF power amplifier operates in the first operation mode for thesaturation type nonlinear amplification, the overcurrent detectioncircuit performs overcurrent protection of the final stage transistor.9. An RF power amplifier apparatus comprising: a first RF poweramplifier; and a second RF power amplifier, wherein the first RF poweramplifier amplifies a first RF transmit input signal having a firstfrequency band ranging from approximately 0.8 GHz to approximately 1.0GHz, wherein the second RF power amplifier amplifies a second RFtransmit input signal having a second frequency band ranging fromapproximately 1.7 GHz to approximately 2.0 GHz, wherein respective poweramplifiers of the first RF power amplifier and the second RF poweramplifier respectively include final-stage amplifier stages each ofwhich amplifies an RF signal to generate an RF transmit output signalsupplied to an antenna of a communication terminal, and signal detectorseach of which detects a level of the RF transmit output signal, whereinthe final-stage amplifier stage of each of the power amplifiers includesa final stage transistor which generates the RF transmit output signalat an output electrode thereof, and a final stage load element coupledbetween the output electrode of the final stage transistor and a sourcevoltage, wherein each of the power amplifiers executes a first operationmode for saturation type nonlinear amplification and a second operationmode for non-saturation type nonlinear amplification, wherein in therespective power amplifiers, the input terminals of the signal detectorsare respectively coupled to the output electrodes of the final stagetransistors to thereby generate RF detection signals each responsive tothe level of the RF transmit output signal from the output terminals ofthe signal detectors, the power amplifiers further including firstdetectors, second detectors and control circuits respectively, whereinin the respective power amplifiers, input terminals of the firstdetectors are respectively supplied with the RF detection signals fromthe output terminals of the signal detectors to thereby generate firstdetection signals from the output terminals of the first detectors,wherein in the respective power amplifiers, input terminals of thesecond detectors are respectively coupled to the output electrodes ofthe final stage transistors to thereby generate second detection signalsfrom the output terminals of the second detectors, wherein in each ofthe power amplifiers, the first detection signal generated from theoutput terminal of the first detector contains a first detectioncomponent responsive to a variation in the level of the RF transmitoutput signal due to a load variation of the antenna when the poweramplifier operates in the first operation mode for the saturation typenonlinear amplification, wherein in each of the power amplifiers, thesecond detection signal generated from the output terminal of the seconddetector contains a second detection component responsive to an increasein the output voltage of the output electrode of the final stagetransistor due to an overload state of the antenna when the poweramplifier operates in the second operation mode for the non-saturationtype linear amplification, wherein in the power amplifiers, the firstand second detectors have first and second input threshold voltagesrespectively, wherein a level of the second input threshold voltage isset higher than that of the first input threshold voltage, wherein ineach of the power amplifiers, the first detection signal generated fromthe output terminal of the first detector and the second detectionsignal generated from the output terminal of the second detector aresupplied to the control circuit, wherein the control circuit controlsthe levels of the RF transmit output signal and the output voltage atthe output electrode of the final stage transistor, wherein in each ofthe power amplifiers, first feedback control by the first detector andthe control circuit reduces the variation in the RF transmit outputsignal generated from the output electrode of the final stage transistorresponsive to the load variation of the antenna upon the operation inthe first operation mode, and wherein in each of the power amplifiers,second feedback control by the second detector and the control circuitreduces the increase in the output voltage of the output electrode ofthe final stage transistor responsive to the overload state of theantenna upon the operation in the second operation mode.
 10. The RFpower amplifier apparatus according to claim 9, wherein in the firstoperation mode for the saturation type nonlinear amplification, the RFsignal is of a constant envelope signal in GSM communications, andwherein in the second operation mode for the non-saturation type linearamplification, the RF signal is of an envelope change signal in eitherone of EDGE and WCDMA communications.
 11. The RF power amplifierapparatus according to claim 10, wherein in each of the poweramplifiers, the signal detector is a directional coupler having a mainline and a sub-line disposed close to each other approximately inparallel.
 12. The RF power amplifier apparatus according to claim 10,wherein in each of the power amplifiers, an output matching circuit iscoupled to the output electrode of the final stage transistor, whereinthe main line of the directional coupler is coupled between the outputmatching circuit and the antenna, and wherein an RF detection signalresponse to the level of the RF transmit output signal is generated fromthe sub-line of the directional coupler.
 13. The RF power amplifierapparatus according to claim 12, wherein each of the power amplifiersfurther includes a first-stage amplifier stage, wherein in each of thepower amplifiers, the RF signal amplified by the final-stage amplifierstage is generated from the first-stage amplifier stage, wherein in eachof the power amplifiers, the first-stage amplifier stage includes afirst stage transistor which generates the RF signal, wherein aprogrammable gain amplifier for supplying an RF transmit input signal iscoupled to a control input terminal of the first stage transistor, andwherein in each of the power amplifiers, the control circuit controlsprogrammable gain of the programmable gain amplifier to thereby controlthe levels of the RF transmit output signal and the output voltage atthe output electrode of the final stage transistor.
 14. The RF poweramplifier apparatus according to claim 12, wherein the control circuitin each of the power amplifiers includes an adder which adds a componentof the first detection signal generated from the output terminal of thefirst detector and a component of the second detection signal generatedfrom the output terminal of the second detector, and a low-pass filtercoupled to the output of the adder, and wherein in each of the poweramplifiers, an analog control signal of the control circuit whichcontrols the levels of the RF transmit output signal and the outputvoltage at the output electrode of the final stage transistor isgenerated from the output of the low-pass filter.
 15. The RF poweramplifier apparatus according to claim 13, wherein in each of the poweramplifiers, the first stage transistor of the first-stage amplifierstage and the final stage transistor of the final-stage amplifier stageare respectively either one of an LDMOS transistor and a hetero bipolartransistor.
 16. The RF power amplifier apparatus according to claim 11,wherein each of the power amplifiers further includes a current sensecircuit and an overcurrent detection circuit, wherein in each of thepower amplifiers, the current sense circuit includes a current sensetransistor which has an element size smaller than the final stagetransistor of the final-stage amplifier stage and which allows a sensecurrent smaller than a current flowing through the final stagetransistor to flow therethrough, wherein in each of the poweramplifiers, the overcurrent detection circuit supplies an overcurrentdetection signal to the control circuit in response to the sense currentflowing through the current sensor transistor of the current sensecircuit, and wherein when the current flowing through the final stagetransistor is brought to an overcurrent state due to the load variationof the antenna when each of the RF power amplifiers operates in thefirst operation mode for the saturation type nonlinear amplification,the overcurrent detection circuit performs overcurrent protection of thefinal stage transistor in each of the power amplifiers.
 17. An RF poweramplifier comprising: a final-stage amplifier stage which amplifies anRF signal to generate an RF transmit output signal supplied to anantenna of a communication terminal; a control circuit; a current sensecircuit; and an overcurrent detection circuit, wherein the final-stageamplifier stage includes a final stage transistor for generating the RFtransmit output signal at an output electrode thereof, and a final-stageload element coupled between the output electrode of the final stagetransistor and a source voltage, wherein the control circuit controls alevel of the RF transmit output signal of the output electrode of thefinal stage transistor in the final-stage amplifier stage, wherein thefinal-stage amplifier stage executes a first operation mode forsaturation type nonlinear amplification and a second operation mode fornon-saturation type linear amplification, wherein in the first operationmode for the saturation type nonlinear amplification, the RF signal isof a constant envelope signal in GSM communications, wherein in thesecond operation mode for the non-saturation type linear amplification,the RF signal is of an envelope change signal in either one of EDGE andWCDMA communications, wherein the current sense circuit includes acurrent sense transistor which has an element size smaller than thefinal stage transistor of the final-stage amplifier stage and whichallows a sense current smaller than a current flowing through the finalstage transistor to flow therethrough, wherein the overcurrent detectioncircuit supplies an overcurrent detection signal to the control circuitin response to the sense current flowing through the current sensortransistor of the current sense circuit, wherein when the RF poweramplifier operates in the first operation mode for the saturation typenonlinear amplification, the overcurrent detection circuit transmits theRF transmit output signal corresponding to the constant envelope signalin the GSM communications, wherein when the RF power amplifier operatesin the second operation mode for the non-saturation type linearamplification, the overcurrent detection circuit transmits the RFtransmit output signal corresponding to the envelope change signal ineither one of the EDGE and WCDMA communications, and wherein when thecurrent flowing through the final stage transistor is brought to anovercurrent state due to a variation in load at the antenna upontransmission of the RF transmit output signal corresponding to theconstant envelope signal in the GSM communications by the firstoperation mode of the RF power amplifier, the overcurrent detectioncircuit performs overcurrent protection of the final stage transistor.18. The RF power amplifier according to claim 17, further including anoutput matching circuit, a directional coupler, a first detector and asecond detector, wherein the output matching circuit is coupled to theoutput electrode of the final stage transistor of the final-stageamplifier stage, wherein the directional coupler has a main line and asub-line disposed close to each other approximately in parallel, whereinthe main line is coupled between the output matching circuit and theantenna, wherein an RF detection signal response to the level of the RFtransmit output signal is generated from the sub-line, wherein an inputterminal of the first detector is supplied with the RF detection signalsent from the sub-line of the directional coupler to thereby generate afirst detection signal from an output terminal of the first detector,wherein an input terminal of the second detector is coupled to theoutput electrode of the final stage transistor to thereby generate asecond detection signal from an output terminal of the second detector,wherein the first detection signal generated from the output terminal ofthe first detector contains a first detection component responsive to avariation in the level of the RF transmit output signal due to thevariation in load at the antenna when the RF power amplifier operates inthe first operation mode for the saturation type nonlinearamplification, wherein the second detection signal generated from theoutput terminal of the second detector contains a second detectioncomponent responsive to an increase in output voltage (Vds) of theoutput electrode of the final stage transistor due to an overload stateof the antenna when the RF power amplifier operates in the secondoperation mode for the non-saturation type linear amplification, whereinthe first detector and the second detector have a first input thresholdvoltage and a second input threshold voltage respectively, wherein alevel of the second input threshold voltage is set higher than a levelof the first input threshold voltage, wherein the first detection signalgenerated from the output terminal of the first detector and the seconddetection signal generated from the output terminal of the seconddetector are supplied to the control circuit, wherein the controlcircuit controls the levels of the RF transmit output signal and theoutput voltage at the output electrode of the final stage transistor,wherein first feedback control by the first detector and the controlcircuit reduces the variation in the RF transmit output signal generatedfrom the output electrode of the final stage transistor responsive tothe load variation of the antenna upon the operation in the firstoperation mode, and wherein second feedback control by the seconddetector and the control circuit reduces the increase in the outputvoltage of the output electrode of the final stage transistor responsiveto the overload state of the antenna upon the operation in the secondoperation mode.
 19. The RF power amplifier according to claim 18,wherein the first stage transistor of the first-stage amplifier stageand the final stage transistor of the final-stage amplifier stage arerespectively either one of an LDMOS transistor and a hetero bipolartransistor.